Code division multiple access (CDMA) transmission system

ABSTRACT

A large-capacity CDMA transmission system that can realize communication with a moving unit such as an automobile, transmitting more quantity of information than the conventional system without increasing the occupied band width, using the same or narrower frequency band width. 
     This system assumes the code division multiple access (CDMA) transmission system for simultaneous multiple communication, which performs phase modulation on a carrier signal while maintaining a phase of the carrier signal in a predetermined period at a predetermined value, so as to generate a primary modulated wave, and then, multiplies the obtained primary modulated wave by a spread code sequence. On the transmitting side, a differential coding phase modulation (DPSK) is employed to generate a primary modulated wave. On the other hand, on the receiving side, a quasi-synchronous detection and difference operation are utilized to detect the phase difference between the last symbol interval and the current symbol interval, and the detected phase difference is given as the information of the current symbol interval.

TECHNICAL FIELD

The present invention relates to the code division multiple access(CDMA) transmission system for transmitting high-speed digitalinformation, in particular under severe fading environment such as inthe mobile communication.

BACKGROUND ART

Recently, in the field of the mobile communication such as theautomobile telephone and the pedestrian telephone, the code divisionmultiple access (CDMA) transmission system has come into practical use.The principle of the CDMA transmission system will be summarized asfollows.

A carrier signal in each symbol period of a certain length is subjectedto Phase Shift Keying (PSK) according to information to be transmitted,to generate a primary modulated wave. In that case, if necessary,primary modulated waves are generated more than the number n of channelsused for the code division multiple access.

The primary modulated wave is multiplied by.a spread code of a specifiedcode length such as 32 or 64, repeating a plurality of times equal tothe number of segments in each symbol period, to generate a spreadsignal. The length of this spread code sequence is equal to theso-called spread factor. Further, it is assumed that spread codes areprovided by a Walsh function or the like, and orthogonal to one another.Further, in a single spread code in each symbol period, a period fromthe first code to the last code is called a segment. Each symbol periodconsists of a plurality of segments. In particular, a segmentcorresponding to a spread code is called “transmission basic segment” orsimply “basic segment”. As referred to in describing the technique ofdespreading, a segment corresponding to a despread code is similarlycalled “basic segment” or simply “basic segment”.

For every transmission basic segment and reception basic segment, theleading edge point and the trailing edge point coincide respectivelywith the point in the leading edge of the first code and the point inthe trailing edge of the last code of the spread code or despread code.Except for delay times such as a processing delay and a transmissiondelay in transmission and reception, the transmission basic segment andthe reception basic segment coincide temporally with each other. In thissense, “transmission basic segment” and “reception basic segment” aregenerally called “basic segment”, and differentiated from “virtualsegment”. Here, “virtual segment” is a new concept disclosed in thepresent invention for improving receiving performance. Further,“transmission segment” is used for specifying a segment used fordespread starting from a non-coincident point of time.

Adding n spread codes required for multiple access simultaneouslyutilizing n channels and at least one spread code indicating phasecompensation information and control information required for thecommunication system, the sum total obtained by adding at least (n+1)spread signals is transmitted.

On the other hand, on a receiving side, control information such as chiptiming, symbol timing, symbol period, and segment period is detectedfrom a received wave. Then, the received wave is multiplied by adespread code corresponding to the spread code used at the time oftransmission, so that the sum total of the received segments for theperiod during which the despread code continuity is obtained, todetermine the despread value. For each symbol, the despread signal isobtained from the despread values of a plurality of received segmentsexisting within the symbol period, to demodulate the primary modulatedwave from a despreading circuit and to detect information phase withinthe symbol period. The phase value detected in this way is called adetected phase value. In mobile communication, severe fading occursfrequently, so that a phase error is generated in the detected phasevalue, largely deviating from a correct phase value. Thus, in order tocompensate a phase error, a known phase value such as zero is subjectedto the primary modulation and then a pilot signal obtained by spreadingwith a specific spread code, which is transmitted at the same time withthe primary modulation. When the pilot signal is received, it ispossible to know the phase error from the previously-known phase value.Assuming that phase errors of the same value arise with respect to allthe spread codes, and subtracting that phase error from the detectedphase value to correct it, disturbance due to fading etc. can besuppressed.

Next, information corresponding to the detected phase value corrected isidentified, thus accomplishing transmission of the information.

Such conventional technique of CDMA transmission system will be furtherdescribed referring to the drawings. In the figures, the same or likenumerals or symbols denote the same or like components.

FIG. 26 shows an outlined configuration of an ordinary CDMA transmitter.In FIG. 26, a known value used for a pilot signal is inputted through aninput terminal 100 and information values are inputted through ninformation input terminals 101-10 n, to corresponding phase modulationcircuits (MOD) 110 and 111-11 n, respectively. The number n of theinputted information values means the number of channels simultaneouslyutilized in multiple access.

According to the inputted information, the phase modulation circuitsperform phase modulation of a carrier signal to generate (n+1) primarymodulated waves corresponding to the signals received through the inputterminals 100-10 n, respectively.

Spreading circuits (SS) 120-12 n obtain the products of correspondingprimary modulated waves and the spread codes applied from spread codegenerating circuits (CG) 130-130 n, synchronously with thecorrespondence of the spread codes with the period of time (chipperiod), and output the obtained products as the spread codes,respectively. Here, the spread codes generated by the spread codegenerating circuits (CG) 130-13 n are orthogonal to one another.Further, the spread code generating circuits (CG) are synchronous withone another, and generate spread codes corresponding to each line of aWalsh function and having the code length N more than or equal to (n+1),within one symbol period and repeating a plurality of timescorresponding to the number of the segments, respectively.

Then, (n+1) spread codes and various control signals are summed in asumming circuit (SUM) 140. Output of the summing circuit (SUM) 140 islimited in its frequency band width by a bandlimiting circuit (BPF) 141,and if necessary, subjected to frequency conversion and poweramplification in a transmitting circuit (TX) 142, prior to transmission.

Now, operation of the phase modulation circuits (MOD) 110-11 n in theabove-mentioned FIG. 26 will be described in detail in the following.Namely, in each of the phase modulation circuits (MOD) 110-11 n, thecarrier signal is divided into periods of a prescribed period T as shownby the primary modulated wave and symbol structure of FIG. 27. Phase ofthe carrier signal is modulated so that a phase of each periodcorresponds one-to-one to a symbol value 00, 01, 10, or 11 transmittedin one period, in accordance with the bit arrangement of QPSK shown inFIG. 28 or the bit arrangement of π/4-shifted QPSK shown in FIG. 29, togenerate a primary modulated wave.

Here, the primary modulated wave generally refers to phase-modulatedsignals generated by QPSK, offset QPSK, differential QPSK, π/4-shifteddifferential QPSK, or the like. Further, when, as described above, QPSKis used for generating a primary modulated wave, it is assumed that aphase of a QPSK wave takes four kinds of values 0, 90, 180 and 270degrees (or, 0, ±90 and ±180 degrees). When π/4-shifted QPSK is used,phase information of a QPSK wave takes four kinds of values 45, 135, 225and 315 degrees (or, ±45 and ±135 degrees). Phase values are residues of360 degrees, and phases of QPSK waves are set to divide the total phasespace into the maximum parts. For example, when it is assumed that thereference phase is 0 degree in QPSK or 45 degrees in π/4-shifted QPSK,all the phases are spaced from each other at intervals of 90 degrees. Bymaking four kinds of phases of primary waves correspond to the states00, 01, 10 and 11, a transmission bit series can be made to correspondto a series of dibits each being a combination of two bits. Thus, eachsymbol can transmit two bits.

As shown in the figures, the symbols are set as 00, 01, 11, and 10counterclockwise in order that a Hamming distance corresponding toadjacent phases becomes 1 and a Hamming distance corresponding tonon-adjacent phases becomes 2. Here, the Hamming distance means thenumber of different bit values between states. For example, distance(00, 01), distance (01, 11), distance (11, 10) and distance (10, 00) areall 1, while distance (00, 11) and distance (10, 01) are each 2.

Such phase-to-state mapping is called Gray coding, and used forsuppressing probability of a transmitted information error due todisturbances during propagation to a lower level. Logically, even whenthe received phase is shifted more than 45 degrees from the transmittedphase due to a disturbance and taken erroneously as an adjacent state,one bit out of the two is saved since the distance from the adjacentstates is always set at 1.

As a matter of course, when phase error of 135 degrees or more arises,all the two bits become errors. In that case, however, any stateassignment leads to all bit error, which can not be saved by Graycoding. Thus, it is impossible to save such an error without introducingan error correcting code or the like.

On the other hand, the symbol period T is a quantity defined by thereciprocal of the symbol rate. For example, when the symbol rate is 32 ksymbol/sec. (hereinafter, symbol/sec will be expressed as sps), Tbecomes T=31.25 μseconds. When, the symbol rate is 32 ksps, thetransmission speed of QPSK becomes 64 k bit/sec. (hereinafter, bit/sec.will be expressed as bps).

Next, operation of the spreading circuits (SS) 120-12 n of FIG. 26 willbe described in more detail.

Now, as shown in FIG. 30, each symbol period of a primary modulated waveis divided into four segment intervals, Segment 0 through Segment 3,each being the same period of time. Here, is given a description of thecase of four segments per symbol. However, the other cases are similarand can be understood by analogy. Accordingly, their description will beomitted. Further, as shown in FIG. 31, each segment interval is dividedinto chip intervals, the number of which is equal to the number of codesin the spread code sequence. Further, it is assumed that a chip value isgiven by the product of a primary modulated wave and a spread code valuein each chip interval. Since a primary modulated wave is a function oftime, time resolution of a chip value is the chip period τ. However,since CDMA performs spreading operation and later-described despreadingoperation on a receiving side, time resolution of transmittedinformation is the segment period τN. Here, N is the code length.

The waveform of the spread signal shown in FIG. 31 shows a case in whichthe first code of a Walsh function having code length of 32 is used asthe spread code.

Generally, a Walsh function is given by the following recurrenceformula. $\begin{matrix}{W_{2N} = {\begin{matrix}W_{N} & W_{N} \\W_{N} & \overset{\_}{W_{N}}\end{matrix}}} & (1)\end{matrix}$where W_(2N) is a square matrix of 2N×2N,

W_(N) is a square matrix of N×N, and

{overscore (W_(N))} is a square matrix of N×N whose elements arecomplements of the elements of W_(N).

For example, W₂ whose element is square matrix of 1×1, i.e. a scalar, isgiven as follows. $\begin{matrix}{W_{2} = {\begin{matrix}0 & 0 \\0 & 1\end{matrix}}} & (2)\end{matrix}$

A row of such Walsh function W_(2N) is used as a code sequence. Here,however, 0 of the Walsh function is made to correspond to “−1” and 1 ofthe Walsh function to “1”. Then, when, for example, the i-th row issynchronized with chip times, zeroth to 31st colums, within a segment, acode sequence called an i-th Walsh code sequence is obtained. Here,0≦i≦N−1, and N is the rank of the Walsh function.

Although it is not necessary to use a Walsh code as a spread code, it isnecessary that the code sequences are orthogonal to each other. Here,when the inner product of codes is zero, then it is said that thosecodes are orthogonal. Further, a Walsh code having a code length of 32will be described. However, description of the other cases such as thecode length of 64 will be omitted, since they are similar, and can beeasily understood by analogy.

Here, orthogonality will be briefly examined taking two or threeexamples of Walsh codes.

Zeroth through second Walsh code sequences are respectively given asfollows.

-   -   0th code: {−1, −1, −1, −1, . . . , −1, −1, −1, −1}    -   1st code: {−1, 1, −1, 1, . . . , −1, 1, −1, 1}    -   2nd code: {−1, −1, 1, 1, . . . , −1, −1, 1, 1}

The inner product of the 0th and 1st codes, inner product {0, 1}, can becalculated as follows.inner product {0, 1}=1−1+1−1+ . . . +1−1+1−1=0

Similarly, the inner product of the 1st and 2nd Walsh codes, innerproduct {1, 2}, and the inner product of the 0th and 2nd codes, innerproduct {0, 2} can be calculated respectively as follows.inner product {1, 2}=1−1−1+1+ . . . +1−1−1+1=0inner product {0, 2}=1+1−1−1+ . . . +1+1−1−1=0

Since all these inner products are 0, it is clear that the codes of theWalsh function are orthogonal to one another. The other cases can beeasily examined, and description of them is omitted.

On the other hand, inner products of the Walsh codes themselves can becalculated as follows. Namely,inner product {0, 0}=1+1+1+1+ . . . +1+1+1+1=32inner product {1, 1}=1+1+1+1+ . . . +1+1+1+1=32inner product {2, 2}=1+1+1+1+ . . . +1+1+1+1=32

When normalized by the code length 32, the inner products of all thecode themselves become always unit 1. When Walsh codes are used as codesequences, this means that the same Walsh code can be used as a spreadcode and the despread code.

Now, it is assumed that, in a certain segment period, theabove-mentioned 0th-2nd Walsh codes are used to transmit three pieces ofinformation by multiplex transmission. When the 0th Walsh code is usedto transmit a value a, the 1st Walsh code to transmit a value b, and the2nd Walsh code to transmit a value c, then information inputted to thesumming circuit (SUM) 140 (summed signal {0, 1, 2}) is describedcorrespondingly to the chips as follows.summed signal {0, 1, 2 }=+a{−1, −1, −1, −1, . . . , −1, −1, −1, −1}+b{−1, 1, −1, 1, . . . , −1, 1, −1, 1}+c{−1, −1, 1, 1, . . . , −1, −1, 1, 1}={−a−b−c, −a+b−c, −a−b+c, −a+b+c, . . . , −a−b−c, −a+b−c, −a−b+c, −a+b+c}  (3)

When the receiving side can correctly receive the summed signal, thereceived summed signal is multiplied by the despread code, to obtain thevalue of the primary modulated signal in the corresponding segment, asfollows.

Namely, the value corresponding to the 0th Walsh code sequence is givenby the inner product of the summed signal {0, 1, 2} and the 0th Walshsignal, as follows. Namely,summed signal {0, 1, 2}-0th Walsh code=−(−a−b−c)−(−a+b−c)−(−a−b+c)−(−a+b+c)−(−a−b−c)−(−a+b−c)−(−a−b+c)−(−a+b+c)=+a+b+c+a−b+c+a+b−c+a−b−c+a+b+c+a−b+c+a+b−c+a−b−c=32a  (4)

Accordingly, it is clear that, when the inner product of the summedsignal {0, 1, 2} and the 0th Walsh code is normalized by the code length32, the value a is correctly received while the values b and c arecompletely suppressed, realizing correct receiving without interference.

Further, the value corresponding to the 1st Walsh code is given by theinner product of the summed signal {0, 1, 2}and the 1st Walsh code asfollows. Namely,summed signal {0, 1, 2}-1st Walsh signal=−(−a−b−c)+(−a+b−c)−(−a−b+c)+(−a+b+c)−(−a−b−c)+(−a+b−c)−(−a−b+c)+(−a+b+c)=+a+b+c−a+b−c+a+b−c−a+b+c+a+b+c−a+b−c+a+b−c−a+b+c=32b  (5)

Accordingly, it is clear that, when the inner product of the summedsignal {0, 1, 2} and the 1st Walsh code is normalized by the code length32, the value b is correctly received, while the values a and c arecompletely suppressed.

Still further, the value corresponding to the 2nd Walsh code is given bythe inner product of the summed signal {0, 1, 2} and the 2nd Walsh codeas follows. Namely,summed signal {0, 1, 2}-2nd Walsh code=−(−a−b−c)−(−a+b−c)+(−a−b+c)+(−a+b+c)−(−a−b−c)−(−a+b−c)+(−a−b+c)+(−a+b+c)=+a+b+c+a−b+c−a−b+c−a+b+c+a+b+c+a−b+c−a−b+c−a+b+c=32c  (6)

Accordingly, it is clear that, when the inner product of the summedsignal {0, 1, 2} and the 2nd Walsh code is normalized by the code length32, the value c is correctly received, while the values a and b arecompletely suppressed.

Thus, as long as spread codes are orthogonal to one another, multipleaccess in which the number of active channels is same as the number ofthe spread codes is possible, and communication can be conducted onlywhen the spread codes on both sides of the communication coincide witheach other. That is the reason that the Code Division Multiple Accesstransmission system can be realized using a code, which is the thirdaxis orthogonal to both the time and frequency axes, as a key forcommunication, in contrast with the Frequency Division Multiple Accesstransmission system in which a carrier frequency is used as a key andthe Time Division Multiple Access transmission system in which a timeslot is used as a key. Further, in CDMA, since it can be considered thata code determines a transmission path, a channel is established for eachspread code. Thus, frequently, the number of the spread codes is calledthe number of channels.

In FIG. 31, since the primary modulated wave has positive values 0-1 inthe 0th segment of the symbol 1, and negative values 0-−1 in the 1stsegment as shown in FIG. 30, the sign of the corresponding chip valuesin the 0th segment of the symbol 1 changes from minus to plusalternately and the sign of the chip values in the 1st segment changesfrom plus to minus alternately.

When the code length is 32, the chip rate of the spread codes becomes 32ksps-4 segments-32 chip/segment=4.096 M chips/sec. (hereinafter,chips/sec. is written as cps).

All the spread codes changes synchronously with one another in each chipinterval, and thus the summed signal whose signal value in a chipinterval is the summation of the hip values becomes a rectangular waveof a constant value within a chip interval. Accordingly, both in thecase of the maximum information rate of 2 Mbps in which 32 channels areused for simultaneous transmission and in the case of the minimuminformation rate in which 1 channel is used for transmission at 64 kbps,the chip rate is always constant at 4.096 Mcps irrespective oftransmission rates of information.

Thus, as shown in FIG. 26, a plurality of spread signals correspondingto information signals and necessary control signal are generated by thespreading circuits (SS) 120-12 n, using spread codes outputted from thespread code generating circuits (CG) 130-13 n and orthogonal to oneanother. Then, the summation of the plurality of spread codes isobtained by the summing circuit (SUM) 140 and, if necessary, the summedsignal obtained is subjected to the frequency conversion and poweramplification in the transmitting circuit (TX), to be transmitted as aCDMA signal.

Here, it is assumed that the number (n+1) of the spreading circuits (SS)120-12 n is equal to or less than the spread factor, i.e., the length Nof spread code sequence.

In order to transmit only the sum total of the summed signal, atransmission band of half the chip rate, i.e. 2.048 MHz is sufficientfrom the Shannon's sampling theorem. However, it is necessary to obtainthe inner product of the received chip waveform and the despread codefor despreading, and thus, it difficult to ensure the orthogonalitybetween the spread codes by transmitting only the sum total. Because ofthis, it is desirable to transmit the rectangular waveform of the summedsignal as faithfully as possible, and accordingly the band width of2.048 MHz or more is used.

Accurate transmission of a rectangular waveform of a CDMA signalrequires a frequency several times as high as the chip rate. However, asshown in FIG. 26, in many cases, band-pass filter operation is carriedout as a function of the bandlimiting circuit (BPF) 141, to limit thefrequency band width to the degree of the chip rate.

As shown in FIG. 26, if necessary, the CDMA signal is subjected tosuitable processing such as conversion to a target frequency and poweramplification in the transmitting circuit (TX) 142, and thereafter,radiated through an antenna. As the target frequency, a frequency domainof 2 GHz is frequently used, and accordingly, the following discussionis directed to CDMA transmission in this frequency domain of 2 GHz.However, the other frequency band is similar and can be easilyconjectured, and their description is omitted. Further, theabove-mentioned control signal does not directly relate to the presentinvention, and therefore, its further description is omitted.

Here, usually, a radio wave transmitted through the transmitting circuit(TX) 142 as described above is seldom transmitted through an ideal radiowave propagation path. In the mobile communication such as theautomobile telephone and the pedestrian telephone, a transmitter itselfmoves so that the Doppler shift is generated and the carrier frequencydeviates. Or, in many cases, a radio wave is received via a plurality ofpropagation paths. Accordingly, the phase and amplitude of the receivedwave changes with time (which is called the fading phenomenon, and inparticular called the Rayleigh fading in a poor transmission environmentcausing the phase change of uniform distribution and making theamplitude have the Rayleigh distribution). Or, a radio wave is stronglyreflected by building walls etc., and accordingly, it arrives at variouspoints of time through different propagation paths of various lengths.In addition, in many cases, these strongly-reflected waves themselvesare transmitted through multi-ray propagation paths with each arrivalwave suffering the Rayleigh fading phenomenon independently.

On the receiving side, a CDMA receiver comprises main functionalcircuits, for example, for reception, synchronous detection, receptioncontrol, demodulation, despread, phase correction, judgment, etc.

In FIG. 32, the reception control circuit (CNT) 204 detects variouscontrol signals required for control of the receiver, and outputs aplurality of despread code sequences required for receiving. Thesynchronous detection circuit (SYNC) 203 outputs a regenerated carrierwave, a chip synchronizing signal, a segment synchronizing signal, asymbol synchronizing signal, etc. from the received signal.

The demodulator circuit (deMOD) 201 has the structure shown in FIG. 33.In that figure, the received wave applied to an input terminal 2010connected to the receiving circuit (RX) 200 is inputted to themultipliers 2011 and 2012. Here, the demodulator circuit (deMOD) 201 ,which generally utilizes the synchronous detection system, obtains theproduct of the regenerated carrier wave 202 and the received wave by themultiplier 2011, then accumulates the product for each carrier cycle bythe accumulator 2014 to obtain the inner product of each carrier cycle,takes in the obtained inner products into the latch register (REG) 2016to hold them only for their carrier cycle periods, and outputs thevalues held in the latch register (REG) 2016 as in-phase components i(t)of the modulated signal of the primary modulated wave, for respectivecarrier cycle periods. At the same time, the demodulator circuit (deMOD)201 obtains an orthogonal carrier signal by shifting the regeneratedcarrier wave 202 by 90 degrees in phase by the phase shifter 2013, toobtain the product of the orthogonal carrier signal and the receivedwave by the multiplier 2012. Then, the product is accumulated for eachcarrier cycle by the accumulator 2015 to obtain the inner product foreach carrier cycle. The obtained inner products are taken into the latchregister (REG) 2017 to hold them only for their carrier cycle periods.The values held in the latch register (REG) 2017 are outputted asquadrature components q(t) of the modulated signal of the primarymodulated wave, for respective carrier cycle periods. The signal Rinputted to the accumulators 2014, 2015 is an accumulation reset signalinputted from the control terminal 2018 for each carrier cycle. At everytrailing edge of this accumulation reset signal R, the accumulatedvalues of the accumulator 2014, 2015 are reset to zero. Further, thesignal R inputted to the latch registers (REG) 2016, 2017 are theaccumulation reset signal inputted from the control terminal 2018 foreach carrier cycle. At every leading edge of this accumulation signal R,the accumulators 2016, 2017 hold the inputted values.

In FIG. 32, the in-phase components i(t) and quadrature components q(t)of the demodulated signal from the demodulator circuit 201 are inputtedto (n+1) despreading circuits (deSS) 210-21 n. FIG. 34 shows an exampleof these despreading circuits (deSS) 210-21 n. An in-phase componenti(t) and quadrature component q(t) of the demodulated signal areinputted to the input terminals 2100, 2101, respectively. Themultipliers 2102, 2103 obtain the products of the in-phase componenti(t) or quadrature component q(t) of the demodulated signal and the i-thdespread code sequence inputted from the terminal 22 i, in accordancewith the chip synchronizing signal, and obtain the accumulation of theproduct for each segment, in accordance with the segment synchronizingsignal. Here, the i-th despread code sequence means the despread codesequence corresponding to the i-th spread code used on the transmittingside. When the Walsh function is used, the despread code sequence andthe spread code sequence are equal to each other.

Accordingly, in FIG. 32, the corresponding despread codes are inputtedto the respective terminals 220-22 n of the despreading circuits 210-21n. Then, outputs of the multipliers 2102, 2103 are accumulated in theaccumulators 2014, 2015. The accumulation reset signal R is inputted tothe accumulators 2104, 2105 from the terminal 2110, for each segment.The outputs of the accumulators 2104, 2105 are each normalized by thecode length, held by the latch registers (REG) 2106, 2107 for thesegment interval, and outputted from the output terminals 2108, 2109 asthe in-phase component I_(i)′(t) and quadrature component Q_(i)′(t) ofthe despread signal.

Here, since the spread codes are orthogonal to one another, when thedesperad code coincides with the spread code of the transmission,despreading circuits 210-21 n output a finite value, realizing correctreceiving. On the other hand, when the despread code does not coincidewith the spread code of the transmission, the despreading circuits210-21 n always output zero, and thus, does not effectively output thereceived signal.

The in-phase components I_(i)′(t) and quadrature components Q_(i)′(t) ofthe despread signal relating to n information channels of thesimultaneous multiple access system are outputted from the despreadingcircuits 211-21 n. The in-phase component I₀′(t) and quadraturecomponent Q₀′(t) of the despread signal relating to the pilot signalcommon to those n channels are outputted from the despreading circuit210.

These despread signals are each subject to disturbances such as phaseerror, amplitude distortion, delay, and the like, during transmission.By transmitting a pilot spread signal obtained by spreading a primarymodulated wave of phase information of a known value, for example “0”,and by measuring the error between the known value and a phase valuedetected on the receiving side, it is logically possible to generallyknow the phase error due to the disturbances that have arisen during thetransmission. Accordingly; as shown in FIG. 32, in many cases, isemployed the pilot system in which one channel of a pilot signal fortransmitting a known value is added to the n channels of information, togenerally correct disturbances during transmission.

Accordingly, the following description is directed to the case in whichone pilot channel is added to n information channels. However, a case inwhich one pilot channel is added to one information channel and a casein which an in-phase component and quadrature component of a primarymodulated wave in each spread signal are assigned respectively to aninformation channel and a pilot channel are similar and can be easilyunderstood by analogy. Therefore, description of such cases is omitted.

In FIG. 32, outputs of the despreading circuits (deSS) 211-21 n andoutput of the despreading circuit (deSS) 210 are led to the phasecorrection circuits (CMP) 231-23 n. FIG. 35 shows an example ofcofiguration of those phase correction circuits (CMP). The in-phasecomponent I_(i)′(t) and quadrature component Q_(i)′(t) of thedespreading circuit (deSS) 23 i are inputted to the input terminals 2300and 2301, respectively. Further, the in-phase component I_(i)′(t) andquadrature component Q₀′(t) of the despreading circuit 230 are inputtedto the input terminals 2302 and 2303, respectively. Then, the in-phasecomponent I_(i)′(t) of the information channel i is inputted to themultipliers 2310 and 2311, and the quadrature component Q_(i)′(t) of theinformation channel i is inputted to the multipliers 2312 and 2313. Thein-phase component I₀′(t) of the pilot channel is inputted to themultipliers 2310 and 2312, and the quadrature component Q₀′(t) of thepilot channel is inputted to the multipliers 2313 and 2311. The adder2320 outputs the sum of the outputs of the multipliers 2310 and 2313, asthe in-phase component I_(i)(t) of the phase correction signal, to theterminal 2340. Further, the adder 2321 outputs the difference betweenthe output of the multiplier 2312 and the output of the multiplier 2311,as the quadrature component Q_(i)(t) of the phase correction signal, tothe terminal 2341.

In FIG. 32, further, the outputs of the phase correction circuits (CMP)231-23 n are led to the decision circuits (DEC) 241-24 n. When thein-phase component I_(i)(t) and quadrature component Q_(i)(t) of thephase correction signal are inputted, the decision circuits (DEC) 241-24n each obtain a phase angle and a received symbol S_(i)(t) of dibitdefined correspondingly to the received phase angle obtained, and outputit as the corresponding information to the terminal 251-25 n.

Next, a series of processing in the receiver shown in FIG. 32 will bedescribed in detail, using mathematics expressions. The output of thereceiving circuit (RX) 200, i.e. the received signal r(t) is written asthe equation 7. $\begin{matrix}{{r\quad(t)} = {\sum\limits_{j = 1}^{m}\quad{\sum\limits_{i = 0}^{n}\quad{a_{j}\quad(t)\quad W_{i}\left\{ {t - {\delta_{l}\quad(t)}} \right\}\quad{\cos\left\lbrack {{\left\{ {\omega_{c} \pm {\Delta\quad\omega_{j}\quad(t)}} \right\}\left\{ {t - {\delta_{j}\quad(t)}} \right\}} + {\theta_{i}\left\{ {i - {\delta_{j}\quad(t)}} \right\}} + {\Delta\quad\phi_{j}\quad(t)}} \right\rbrack}}}}} & (7)\end{matrix}$

Here, the suffix j indicates j-th propagation path of multi-ray Rayleighfading, when, for the sake of convenience, the propagation paths areexpressed as 1st, 2nd, . . . , m-th propagation paths in order ofaverage received power. The letter m indicates the total number of themulti-ray propagation paths;

The suffix i (i=0, 1, . . . , n) indicates the number of the codesequence of the Walsh function, and here n means the total number of thecode sequences of the Walsh function used for transmission;

δ_(j)(t) indicates a delay time in the propagation path j;

a_(j)(t) indicates an amplitude distortion in the propagation path j. Itis assumed that a_(j)(t) is given by a_(j)(t)=α_(j)(t)k_(j)(t). Here,α_(j)(t) is a fading amplitude distortion in the propagation path j, andit is assumed that the amplitude shows the Rayleigh distribution and themaximum variable frequency is defined by the fading frequency. Further,k_(j)(t) is the propagation gain of the propagation path j;

Δφ_(j)(t) indicates the fading phase error of the propagation path j,and its value is uniformly distributed between −180 degrees and 180degrees. It is assumed that the upper limit of the variable frequency isdefined by the fading frequency;

W_(i)(t) is the value of the i-th spread code sequence at the time t,the i-th spread code sequence changing correspondingly to the chip;

ω_(c) is defined by ω_(c)=2πf_(c), where f_(c), is the carrierfrequency;

Δω_(j)(t) indicates the frequency deviation caused by Doppler shift inthe propagation path j; and

θ_(i)(t) indicates the information phase of the primary modulated wavecorresponding to the i-th code sequence.

Here, in the synchronous detection circuit (SYNC) whose details are notshown, the carrier signal is regenerated in accordance with thecomponents that have passed a plurality of propagation paths andcontained in the received wave. These regenerated carrier in-phase wavec(t) and regenerated carrier quadrature wave s(t) are respectively givenas follows.c(t)=cos [ω_(c) {t−δ(t)}+Δφ(t)]  (8)s(t)=sin [ω_(c) {t−δ(t)}+Δφ(t)]  (9)where δ(t) is a time delay of the regenerated carrier wave, and Δφ(t) isa phase error of the regenerated carrier wave.

The outputs of the demodulator circuit (deMOD) 201 , i.e., the in-phasecomponent i(t) and quadrature component q(t) of the demodulated signalare respectively given as the inner products of the received signal r(t)and the regenerated carrier in-phase wave c(t) or the regeneratedcarrier quadrature wave s(t) as follows. $\begin{matrix}{{i\quad(t)} = {\frac{1}{\tau}\quad{\int_{t}^{1 + \tau}{r\quad(t)\quad c\quad(d)\quad{\mathbb{d}t}}}}} & (10) \\{{q\quad(t)} = {\frac{1}{\tau}\quad{\int_{t}^{1 + \tau}{r\quad(t)\quad s\quad(d)\quad{\mathbb{d}t}}}}} & (11)\end{matrix}$where τ is the carrier cycle period, i.e., the reciprocal of the carrierfrequency.

The carrier cycle is small in comparison with the chip cycle, andfurthermore, the fading cycle and the frequency deviation of the Dopplershift are sufficiently small in comparison with the carrier frequency.Accordingly, it can be assumed that the value of the spread code, thefading phase distortion, and the fading amplitude distortion aremaintained at constant values within a carrier cycle. Here, the fadingcycle means the reciprocal of the fading frequency.

Thus, the equations 10 and 11 can be calculated as follows.$\begin{matrix}{{i\quad(t)} = {\frac{1}{2\quad\tau}\quad{\int_{t}^{1 + \tau}{\sum\limits_{j = 1}^{m}\quad{\sum\limits_{i = 0}^{n}\quad{\alpha_{j}\quad(t)\quad W_{j}\quad{\left\{ {t - {\delta_{j}\quad(t)}} \right\} \cdot \left\lbrack {{\cos\left\{ {{2\quad\omega_{c}\quad t} + {{\theta_{i}\quad\left( {t - {\delta_{j}\quad(t)}} \right)} \pm {\Delta\quad\omega_{j}\quad(t)\quad\left( {t - {\delta_{j}\quad(t)}} \right)}} + {\Delta\quad\phi_{j}\quad(t)} + {\Delta\quad\phi\quad(t)} - {\omega_{c}\quad\left( {{\delta_{j}\quad(t)} + {\delta\quad(t)}} \right)}} \right\}} + {\cos\left\{ {{{\theta_{i}\quad\left( {t - {\delta_{j}\quad(t)}} \right)} \pm {\Delta\quad\omega_{j}\quad(t)\quad\left( {t - {\delta_{j}\quad(t)}} \right)}} + \left( {{\Delta\quad\phi_{j}\quad(t)} - {\Delta\quad\phi\quad(t)}} \right) - {\omega_{c}\quad\left( {{\delta_{j}\quad(t)} + {\delta\quad(t)}} \right)}} \right\}}} \right\rbrack}\quad{\mathbb{d}t}}}}}}} & (12)\end{matrix}$

With respect to the variables for the trigonometric functions within thebrackets [ ], variables other than the component of the carrier arealmost constants within a carrier cycle. Accordingly, with respect tothe in-phase component i(t) of the demodulated signal, the integral ofthe first term within the brackets [ ] converges to zero, and the secondterm becomes almost the average. $\begin{matrix}{{i\quad(t)} = {\frac{1}{2}\quad{\sum\limits_{j = 1}^{m}\quad{\sum\limits_{i = 0}^{n}\quad{\alpha_{j}\quad(t)\quad W_{j}\left\{ {t - {\delta_{j}\quad(t)}} \right\}{\cos\left\lbrack {{\theta_{i}\quad\left( {t - {\delta_{j}\quad(t)}} \right)} + {\varphi_{j}\quad(t)}} \right\rbrack}}}}}} & (13)\end{matrix}$whereφ_(j)(t)=±Δω_(j)(t){t−δ_(j)(t)}+{Δφ_(j)(t)−Δφ(t)}−ω_(c){δ_(j)(t)−δ(t)}  (14)

Similarly, the quadrature component q(t) of the demodulated signal isobtained as follows. $\begin{matrix}{{q(t)} = {{{- \frac{1}{2\quad\tau}}\quad{\int_{t}^{1 + \tau}{\sum\limits_{j = 1}^{m}\quad{\sum\limits_{i = 0}^{n}\quad{\alpha_{j}\quad(t)\quad W_{j}\quad{\left\{ {t - {\delta_{j}\quad(t)}} \right\} \cdot \left\lbrack {{\sin\left\{ {{2\quad\omega_{c}\quad t} + {{\theta_{i}\quad\left( {t - {\delta_{j}\quad(t)}} \right)} \pm {\Delta\quad\omega_{j}\quad(t)\quad\left( {t - {\delta_{j}\quad(t)}} \right)}} + {\Delta\quad\phi_{j}\quad(t)} + {\Delta\quad\phi\quad(t)} - {\omega_{c}\quad\left( {{\delta_{j}\quad(t)} + {\delta\quad(t)}} \right)}} \right\}} + {\sin\left\{ {{{\theta_{i}\quad\left( {t - {\delta_{j}\quad(t)}} \right)} \pm {\Delta\quad\omega_{j}\quad(t)\quad\left( {t - {\delta_{j}\quad(t)}} \right)}} + \left( {{\Delta\quad\phi_{j}\quad(t)} - {\Delta\quad\phi\quad(t)}} \right) - {\omega_{c}\quad\left( {{\delta_{j}\quad(t)} + {\delta\quad(t)}} \right)}} \right\}}} \right\rbrack}\quad{\mathbb{d}t}}}}}} \cong {{- \frac{1}{2}}\quad{\sum\limits_{j = 1}^{m}\quad{\sum\limits_{i = 0}^{n}\quad{\alpha_{j}\quad(t)\quad W_{j}\quad\left( {t - {\delta_{j}\quad(t)}} \right\}\quad{\sin\left\lbrack {{\theta_{i}\quad\left( {t - {\delta_{j}\quad(t)}} \right)} + {\varphi_{j}\quad(t)}} \right\rbrack}}}}}}} & (15)\end{matrix}$

The in-phase component I_(d)′(t) or quadrature component Q_(d)′(t) ofthe despread code of the channel d outputted from the despreadingcircuit (deSS) 210-21 n are given as the inner product between thedespread code sequence W_(d) and the in-phase component i(t) orquadrature component q(t) of the demodulated signal within a segment, asfollows. $\begin{matrix}{{I_{d}^{\prime}\quad(t)} = {\frac{1}{2N}\quad{\sum\limits_{k = 0}^{N - 1}\quad{\sum\limits_{j = 1}^{m}\quad{\sum\limits_{l = 0}^{n}\quad{\alpha_{j}\quad(t)\quad W_{i}\left\{ {t + {k\quad\lambda} - {\delta_{j}\quad(t)}} \right\} W_{d}{\left\{ {t + {k\quad\lambda} - {\delta_{j}\quad(t)}} \right\} \cdot {\cos\left\lbrack {{\theta_{i}\quad\left( {t + {k\quad\lambda} - {\delta_{j}\quad(t)}} \right)} + {\varphi_{j}\quad\left( {t + {k\quad\lambda}} \right)}} \right\rbrack}}}}}}}} & (16)\end{matrix}$ $\begin{matrix}{{Q_{d}^{\prime}\quad(t)} = {\frac{1}{2N}\quad{\sum\limits_{k = 0}^{N - 1}\quad{\sum\limits_{j = 1}^{m}\quad{\sum\limits_{i = 0}^{n}\quad{\alpha_{j}\quad(t)\quad W_{i}\left\{ {t + {k\quad\lambda} - {\delta_{j}\quad(t)}} \right\} W_{d}{\left\{ {t + {k\quad\lambda} - {\delta_{j}\quad(t)}} \right\} \cdot {\sin\left\lbrack {{\theta_{i}\quad\left( {t + {k\quad\lambda} - {\delta_{j}\quad(t)}} \right)} + {\varphi_{j}\quad\left( {t + {k\quad\lambda}} \right)}} \right\rbrack}}}}}}}} & (17)\end{matrix}$where 0≦d≦n, and λ is the chip cycle and N is the code length.φ_(j)(t+kλ)=±Δω_(j)(t){t+kλ−δ_(j)(t)}+{Δφ_(j)(t)−Δφ(t)}−ω{δj(t)−δ(t)}  (18)

When, in the equations 16 and 17, the despread code W_(d) of thereception channel d correctly coincides the transmission code W_(i),then, the in-phase component I_(i)′(t) and quadrature componentQ_(i)′(t) of the despread signal corresponding to the i-th spread codeare respectively given as follows. $\begin{matrix}{{I_{i}^{\prime}\quad(t)} \cong {\sum\limits_{j = 1}^{m}\quad{\frac{{\overset{\sim}{a}}_{j}\quad(t)}{2}\quad\cos\left\{ {{\theta_{i}\quad\left( {t - {\delta_{j}\quad(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}\quad(t)} + {{\overset{\sim}{\varphi}}_{j}\quad(t)}} \right\}}}} & (19) \\{{Q_{i}^{\prime}\quad(t)} \cong {\sum\limits_{j = 1}^{m}\quad{\frac{{\overset{\sim}{a}}_{j}\quad(t)}{2}\quad\sin\left\{ {{\theta_{i}\quad\left( {t - {\delta_{j}\quad(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}\quad(t)} + {{\overset{\sim}{\varphi}}_{j}\quad(t)}} \right\}}}} & (20)\end{matrix}$where

-   ã_(j)(t) is an expected value, within a segment, of the amplitude    distortion a_(j)(t) in the j-th propagation path;-   {tilde over (ψ)}_(i)(t) is an expected value, within a segment, of    the phase error φ_(i)(t) having the frequency characteristics    intrinsic to the spread code sequence W_(i);-   {tilde over (φ)}_(j)(t) is an expected value, within a segment, of    the phase error φ_(j)(t) in the j-th propagation path,    {tilde over (φ)}_(j)(t)=±Δ{tilde over (ω)}_(j)(t){t−{tilde over (δ)}    _(j)(t)}+{Δ{tilde over (φ)}_(j)(t)−Δ{tilde over    (φ)}(t)}−ω_(c){{tilde over (δ)}_(j)(t)−{tilde over (δ)}(t)}  (21)    and    ã_(j)(t)=ã_(j)(t){tilde over (k)} _(j)(t)

ã_(j)(t) is an expected value, within a segment, of the fading amplitudedistortion in the propagation path j;

{tilde over (k)}_(j)(t) is an expected value, within a segment, of thepropagation gain in the propagation path j;

Δ{tilde over (ω)}_(j)(t) is an expected value, within a segment, of theDoppler shift in the propagation path j;

Δ{tilde over (φ)}_(j)(t) is an expected value, within a segment, of thefading phase error in the propagation path j;

{tilde over (δ)}_(j)(t) is an expected value, within a segment, of thepropagation delay in the propagation path j;

Δ{tilde over (φ)}(t) is an expected value, within a segment, of thephase error of the regenerated carrier wave; and

{tilde over (δ)}(t) is an expected value, within a segment, of the delayof the regenerated carrier wave.

The spread code sequence W_(i) of the received wave coming through aninferior propagation path has already been subjected to distortion, andthus, an error φ_(i)(t) intrinsic to the spread code sequence W_(i) isgenerated in the despread signal. Further, the in-phase componentI_(i)′(t) and quadrature component Q_(i)′(t) of the despread signal forthe channel i in the two-ray Rayleigh fading environment are given bythe following simple equation.I _(i)′(t)≅{tilde over (β)}(t)cos {θ_(i)(t−{tilde over (δ)} ₁(t))+{tildeover (ψ)}_(i)(t)+{tilde over (ψ)}(t)}  (22)Q _(i)′(t)≅−{tilde over (β)}(t)sin {θ_(i)(t−{tilde over (δ)}₁(t))+{tilde over (ψ)}_(i)(t)+{tilde over (ψ)}(t)}  (23)where $\begin{matrix}{{\overset{\sim}{\beta}\quad(t)} = {\frac{1}{2}\quad\sqrt{{{\overset{\sim}{a}}_{1}\quad(t)^{2}} + {{\overset{\sim}{a}}_{2}\quad(t)^{2}} + {2\quad{\overset{\sim}{a}}_{1}\quad(t)\quad{\overset{\sim}{a}}_{2}\quad(t)\quad\cos\left\{ {{{\overset{\sim}{\varphi}}_{1}\quad(t)} - {{\overset{\sim}{\varphi}}_{2}\quad(t)}} \right\}}}}} & (24) \\{{\overset{\sim}{\vartheta}\quad(t)} = {\tan^{- 1}\left\lbrack \frac{{{\overset{\sim}{a}}_{1}\quad(t)\quad\sin\left\{ {{\overset{\sim}{\varphi}}_{1}\quad(t)} \right\}} + {{\overset{\sim}{a}}_{2}\quad(t)\quad\sin\left\{ {{\overset{\sim}{\varphi}}_{2}\quad(t)} \right\}}}{{{\overset{\sim}{a}}_{1}\quad(t)\quad\cos\left\{ {{\overset{\sim}{\varphi}}_{1}\quad(t)} \right\}} + {{\overset{\sim}{a}}_{2}\quad(t)\quad\cos\left\{ {{\overset{\sim}{\varphi}}_{2}\quad(t)} \right\}}} \right\rbrack}} & (25)\end{matrix}$

Further, the equation 24 can be expressed using the 2nd propagation pathto the 1st propagation path ratio P₂₁(t) of the instantaneous power toobtain the following equation. $\begin{matrix}{{\overset{\sim}{\beta}\quad(t)} = {\frac{{\overset{\sim}{a}}_{1}\quad(t)}{2}\quad\sqrt{1 + {P_{21}^{2}\quad(t)} + {2P_{21}\quad(t)\quad\cos\left\{ {{{\overset{\sim}{\varphi}}_{1}\quad(t)} - {{\overset{\sim}{\varphi}}_{2}\quad(t)}} \right\}}}}} & (26)\end{matrix}$where the instantaneous power ratio P₂₁(t) is defined by${P_{21}\quad(t)} = \frac{{\overset{\sim}{a}}_{2}\quad(t)}{{\overset{\sim}{a}}_{1}\quad(t)}$

Similarly, the equation 25 can be expressed as follows, using theinstantaneous power ratio on its right side. $\begin{matrix}{{\overset{\sim}{\vartheta}\quad(t)} = {\tan^{- 1}\left\lbrack {\frac{\sin\left\{ {{\overset{\sim}{\varphi}}_{1}\quad(t)} \right\}}{\cos\left\{ {{\overset{\sim}{\varphi}}_{2}\quad(t)} \right\}}\quad\frac{1 + {P_{21}\quad(t)\quad\frac{\sin\left\{ {{\overset{\sim}{\varphi}}_{2}\quad(t)} \right\}}{\sin\left\{ {{\overset{\sim}{\varphi}}_{1}\quad(t)} \right\}}}}{1 + {P_{21}\quad(t)\quad\frac{\cos\left\{ {{\overset{\sim}{\varphi}}_{2}\quad(t)} \right\}}{\cos\left\{ {{\overset{\sim}{\varphi}}_{1}\quad(t)} \right\}}}}} \right\rbrack}} & (27)\end{matrix}$

The incoming wave of the 1st propagation path is called a desired wave(D wave) and an incoming wave of a propagation path other than the 1stpropagation path is called an undesired wave (U wave), and their powerratio${P_{12}\quad(t)} = \frac{{\overset{\sim}{a}}_{1}\quad(t)}{{\overset{\sim}{a}}_{2}\quad(t)}$is, in particular, defined as the instantaneous DUR. This instantaneousDUR is the reciprocal in relation to the above-defined instantaneouspower ratio P₂₁(t).

Further, in many times, DUR is defined as the ratio of the time-averageof the power of the D wave to the time-average of the power of the Uwave, and expressed by D/U as a true value or by 10·log₁₀(D/U) as adecibel.

The spread code W_(i) generates the spread signal exhibiting anintrinsic spectrum distribution, and thus, in the frequency-selectivefading environment in which the propagation path itself has thefrequency characteristic, the error {tilde over (ψ)}_(i)(t) shown in theequations 19, 20, 22, and 23 appears strongly.

When the known phase value of the pilot channel is 0 and the channel isassigned to the 0th channel, the in-phase component I₀′(t) andquadrature component Q₀′(t) of the despread signal in the pilot channelare given as follows.I ₀′(t)≅{tilde over (β)}(t)cos {{tilde over (ψ)}₀(t)+{tilde over(ψ)}(t)  (28)Q ₀′(t)≅{tilde over (β)}(t)sin {{tilde over (ψ)}₀(t)+{tilde over(ψ)}(t)  (29)

The phase correction circuit 23 i conducts phase correction shown in thefollowing, to output the in-phase component I_(i)(t) and quadraturecomponent Q_(i)(t) of the correction signal. Namely,I _(i)(t)=I _(i)′(t)I ₀′(t)+Q _(i)′(t)Q ₀′(t)  (30)Q _(i)(t)=Q _(i)′(t)I ₀′(t)−I _(i)′(t)Q ₀′(t)  (31)

Substituting the equations 22, 23, 28, and 29 expressing respectivecomponents of the despread signal into the equations 30 and 31, thecorrection signal is given as follows. $\begin{matrix}\begin{matrix}{{I_{i}(t)} = {{{{\overset{\sim}{\beta}}^{2}(t)}\cos\left\{ {{\theta_{i}\left( {t - {\delta_{1}(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}(t)} + {\overset{\sim}{\vartheta}(t)}} \right\}\cos\left\{ {{{\overset{\sim}{\psi}}_{o}(t)} + {\overset{\sim}{\vartheta}(t)}} \right\}} +}} \\{{{\overset{\sim}{\beta}}^{2}(t)}\sin\left\{ {{\theta_{i}\left( {t - {\delta_{1}(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}(t)} + {\overset{\sim}{\vartheta}(t)}} \right\}\sin\left\{ {{{\overset{\sim}{\psi}}_{o}(t)} + {\overset{\sim}{\vartheta}(t)}} \right\}} \\{= {{{\overset{\sim}{\beta}}^{2}(t)}\cos\left\{ {{\theta_{i}\left( {t - {\delta_{1}(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}(t)} + {{\overset{\sim}{\psi}}_{o}(t)}} \right\}}}\end{matrix} & (32)\end{matrix}$ $\begin{matrix}\begin{matrix}{{Q_{t}\quad(t)} = {{{- {\overset{\sim}{\beta}}^{2}}\quad(t)\quad\sin\left\{ {{\theta_{i}\quad\left( {t - {\delta_{1}\quad(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}\quad(t)} + {\overset{\sim}{\vartheta}\quad(t)}} \right\}\quad\cos\left\{ {{{\overset{\sim}{\psi}}_{0}\quad(t)} + {\overset{\sim}{\vartheta}\quad(t)}} \right\}} +}} \\{{\overset{\sim}{\beta}}^{2}\quad(t)\quad\cos\left\{ {{\theta_{i}\quad\left( {t - {\delta_{1}\quad(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}\quad(t)} + {\overset{\sim}{\vartheta}\quad(t)}} \right\}\quad\sin\left\{ {{{\overset{\sim}{\psi}}_{0}\quad(t)} + {\overset{\sim}{\vartheta}\quad(t)}} \right\}} \\{= {{- {\overset{\sim}{\beta}}^{2}}\quad(t)\quad\sin\left\{ {{\theta_{i}\quad\left( {t - {\delta_{1}\quad(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}\quad(t)} - {{\overset{\sim}{\psi}}_{0}\quad(t)}} \right\}}}\end{matrix} & (33)\end{matrix}$

Using the in-phase component I_(i)(t) and quadrature component Q_(i)(t)of the correction signal in the decision circuits 241-24 n, theinformation phase of the channel i is obtained as follows. And, based onthe obtained information phase of the channel i, the received symbol,i.e., received information of the channel i is decided in accordancewith the rule assigned on the transmission side. The information phaseof the channel i is given as follows.$\begin{matrix}\begin{matrix}{{{information}\quad{phase}_{i}\quad(t)} = {- {\tan^{- 1}\left\lbrack \frac{Q_{i}\quad(t)}{I_{i}\quad(t)} \right\rbrack}}} \\{= {\tan^{- 1}\left\lbrack \frac{\sin\left\{ {{\theta_{i}\quad\left( {t - {\delta_{1}\quad(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}\quad(t)} - {{\overset{\sim}{\psi}}_{0}\quad(t)}} \right\}}{\cos\left\{ {{\theta_{i}\quad\left( {t - {\delta_{1}\quad(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}\quad(t)} - {{\overset{\sim}{\psi}}_{0}\quad(t)}} \right\}} \right\rbrack}} \\{= {{\theta_{i}\quad\left( {t - {\delta_{i}\quad(t)}} \right)} + {{\overset{\sim}{\psi}}_{i}\quad(t)} - {{\overset{\sim}{\psi}}_{0}\quad(t)}}}\end{matrix} & (34)\end{matrix}$

In the last right side of the equation 34 expressing the receptioninformation of the channel i, the first term is a true value of thereceived phase, and the second and subsequent terms indicatedisturbances. The Doppler shift error, the fading phase error, and thedelay error appearing in the equations 12 and 13 now disappear, thusshowing that the phase correction circuits operate effectively. However,it is clear that, as shown by the second and third terms, thefrequency-selective fading errors can not be removed, and remain asfactors deteriorating the communication quality.

The received information may be decided from the information phasewithin a single segment shown in the equation 34. However, the receivedinformation can be decided from the average value of the informationphases in a plurality of segments in the same symbol period, in order toimprove the noise immunity and communication quality.

For example, for obtaining the averages of the in-phase componentI_(i)(t) and quadrature component Q_(i)(t) of the correction signalwithin a symbol, it is noted that the amplitude distortion in a symbolis nearly constant, to obtain the average values using the equations 32and 33. Then, by deciding the information phase from the obtainedaverage values, the communication quality can be further improved.

Further, the averages may be obtained after removing the amplitudedistortions in the equations 32 and 33. Namely, utilizing the fact thatthe amplitude value of the correction signal is obtained by squaring thesum of the square of the in-phase component and the square of thequadrature component, the amplitude distortion can be easily removed.

DISCLOSURE OF THE INVENTION

In order to quantitatively examine effects of the frequency-selectivefading on CDMA in communication of various moving modes in a usual cityarea, computer simulations are conducted as follows.

It is said that typically DUR=25 dB and delay is 1 μsecond, from actualmeasurements of propagation in a city area. However, these valueslargely fluctuate depending on city environment such as buildingheights, wall materials, and road widths, as well as weather conditions.Of course, in conducting the simulations with setting of the followingsystem conditions and moving modes, it is assumed that components of thereceived waves coming through various propagation paths are each subjectto independent Rayleigh fading, arriving in a changing propagation timedepending on city environment.

In these simulations, severe conditions are set in order to make thosesimulations match any real propagation. Namely, as the systemconditions, it is assumed that the transmission frequency domain is the2 GHz band, the chip rate is 4.096 Mcps, the number of the simultaneousmultiple accesses is 31 (at the transmission speed of 1.984 Mbps), thesymbol rate is 32 ksps, and the spread code length and despread codelength are each 32.

For evaluating a system, is used an Eb/No value that gives BER=0.001,i.e. the value generally used in mobile communication in which acompensation function is not used. Here, BER is a ratio of the receivederror bits to the total received bits, and Eb/No is a decibel value dBof a ratio of the received power to the received noise power on thereceiving side. Further, in order to correctly evaluate the CDMA system,the compensation functions such as an error correction coding, RAKEreception, transmission power control, AGC, and the like, which havebeen conventionally and frequently used in the CDMA system, are not usedin the simulations.

When the internal noise in the receiver is sufficiently small, the noiseentering in the course of propagation, i.e. city noise becomes thegoverning term in the received noise power. City noise is independent ofthe communication system, and has a nearly constant value. Thus, thesmaller Eb/No required for realizing a certain BER value is, the smallerthe transmission power corresponding to the gain of Eb/No is, meaningthe communication system is superior.

With respect to environmental conditions, it is insufficient to relyonly on measured values in city areas. And, it is necessary to conductsimulations assuming more severe conditions than the measured values. Asmore severe conditions, is assumed two-ray Rayleigh fading in whichDUR=10 dB, and delay is 1 μsecond. Since, in mobile communication,moving speed of a mobile unit largely affects the communicationcharacteristics, simulations are conducted in the following threetelephone modes.

[Automobile Telephone Mode]

When CDMA conducts communication in the 2 GHz domain, and in fadingenvironment such as a city area, a quasi-constant wave appears near theearth surface, and the wave length of this quasi-constant wave is about0.15 m. When communication is conducted during high-speed travelling of100 km/h along, for example, an express-highway of a city area, then,the maximum Doppler shift is 0.1 ppm (i.e., the maximum frequencydeviation according to the Doppler shift is 200 Hz), and the maximumfading frequency f_(d) is about 185 Hz. Since the symbol rate of CDMA is32 ksps, we obtains the following.f_(d)T≅0.005[Pedestrian Telephone Mode]

When CDMA conducts communication in the 2 GHz domain, and a mobile unitconducts communication in walking at 10 km/h in a city area, the maximumDoppler shift becomes 0.01 ppm and the maximum fading frequency f_(d)becomes about 18.5 Hz. Since the symbol rate of CDMA is 32 ksps, thefollowing result is obtained.f_(d)T≅0.0005[Stationary Telephone Mode]

When CDMA conducts communication in the 2 GHz domain, fading frequencyf_(d) does not completely become zero in a stationary state at a stop ofwalking or driving, although the Doppler shift becomes zero. Asdescribed above, fading is generated as a result of synthesizing manyarrived radio waves that have subjected to reflection, delay, anddiffraction at many places. Also, according to change of physicalconditions, such as a state of air temperature and humiditydistribution, constituting the radio wave propagation path,characteristics affecting radio wave propagation change.

Accordingly, even when a mobile unit is stationary, the propagation pathfluctuates and slow fading appears. As a result, the fading frequency ofthe stationary telephone mode becomes about a tenth of the pedestriantelephone mode. Thus, f_(d) becomes about 1.85 Hz, and the followingresult is obtained.f_(d)T≅0.00005

FIGS. 36, 37 and 38 show respective simulation effects when, withrespect to CDMA of the conventional pilot system, the transmission bandwidth is employed as a parameter and communication is conductedaccording to the above-described three telephone modes. The vertical andhorizontal axes of these figures indicate a bit error rate BER and areceived electric field level Eb/No, respectively.

In the stationary telephone mode (is95.sty) of FIG. 36, high qualitycommunication of BER=0.001 can be realized in all the transmission bandwidths 1.51-25.60 MHz, and in the received electric field area Eb/No≦0dB.

With respect to the explanation IS95.STY. 2.14-is95.STY. 25.60 in thefigure, IS95 means the system name of the conventional CDMA transmissionsystem, STY means the telephone mode of the stationary telephone mode,and 2.14-25.60 are values of the transmission band widths given in MHz.

In the explanation in a series of figures, FIGS. 36-38, 16-18, and20-25, a character string A.B.C is used in a similar manner. Namely,${A.B.C} = {\begin{Bmatrix}{IS95} \\{diffCDMA} \\{IS95\_ CP} \\{diffCDMA\_ CP} \\{diffCDMA\_ CS} \\{ID95\_ CPS} \\{diffCDMA\_ CPS} \\{diffCDMA\_ VSI}\end{Bmatrix} \cdot \begin{Bmatrix}{STY} \\{MAN} \\{CAR}\end{Bmatrix} \cdot \begin{Bmatrix}0.600 \\\vdots \\3.20 \\\vdots \\26.60\end{Bmatrix}}$means a system name, telephone mode, and transmission band width.

With respect to the first string A,

IS95 means the conventional CDMA transmission system, as alreadydescribed;

diffCDMA means the differential CDMA transmission system;

IS95_CP means the conventional CDMA to which the phase continuoustechnique is applied;

diffCDMA_CP means the differential CDMA transmission system to which thephase continuous technique is applied;

diffCDMA_CS means the differential CDMA transmission system to which thechip waveform continuating technique is applied;

IS95_CPS means the conventional CDMA transmission system to which thephase continuous technique and the chip waveform continuating techniqueare applied;

diffCDMA_CPS means the differential CDMA transmission system to whichthe phase continuous technique and the chip waveform continuatingtechnique are applied; and

diffCDMA_VSI means the differential CDMA transmission system to whichthe virtual segment interleave technique is applied.

Further, with respect to the middle string B,

STY means the stationary telephone mode, as already described;

MAN means the pedestrian telephone mode; and

CAR means the automobile telephone mode.

Further, the value of the last string C means the transmission bandwidth given in MHz.

In the pedestrian telephone mode (is95.man) shown in FIG. 37, when thetransmission band width is set at 3.46 MHz or more, it is possible torealize high quality communication with BER=0.001 at the almost samereceiving level as the stationary telephone mode. However, when thetransmission band width is limited to 3.28 MHz or less, there arisefloors at which BER≧0.001, and there appears a defect that high qualitycommunication with BER≦0.001 can not be realized even if the strongesttransmission power is used. Here, using the transmission band width as aparameter, FIG. 37 illustrates, in detail, the neighborhood of the bandwidth at which the floor phenomenon appears. This is because use of theband width value critical to the floor phenomenon is important forquantitatively evaluating the transmission systems. The othertransmission band widths are omitted in the figure, since, for example,the band width of 3.66 MHz or more brings a high quality communicationstate with BER≦0.001 and, on the other hand, the narrower band width of3.20 MHz or less brings a floor. Here, the critical transmission bandwidth is defined by the value of the minimum transmission band widththat realizes high quality communication. In the case of this figure,the critical band width is 3.46 MHz.

In the case of the higher moving speed of the automobile telephone mode(is95.car) as shown in FIG. 38, all the received electric field levelsbring a floor at BER≧0.2 regardless of the transmission band width.Thus, there is a problem that communication is impossible any longer,even with the strongest transmission power and the largest transmissionband width.

Although mobile communication always suffers from the Rayleigh fadingphenomenon, there is no Doppler shift in the stationary telephone modeand the fading phenomenon is not so obvious. Thus, in the stationarytelephone mode, the piloted CDMA can acquire high quality ofcommunication of BER=0.001 even for a transmission path of a rathernarrow band. However, when communication is conducted moving at a slowmoving speed such as about 10 km/h as in the pedestrian telephone modeshown in FIG. 37, the piloted CDMA can provide high quality ofcommunication of BER=0.001 for weak received electric field havingEb/No≦0 dB, similarly to the stationary telephone mode, in the case thatthe transmission band width is 3.66 MHz or more. However, in the casethat the transmission band width is 3.65 MHz or less, a floor isgenerated and it is impossible to conduct communication. Further, whenthe moving speed becomes a high speed of about 100 km/h as in theautomobile telephone mode shown in FIG. 38, there appears a phenomenonthat a floor is generated in the neighborhood of BER=0.2 even for thewidest transmission band width and the highest transmission power,differently from the pedestrian telephone mode, and thus it isimpossible to conduct communication.

Thus, by a hair's breadth, the piloted CDMA can provide a large-capacitycommunication system using sufficient transmission band width, beinglimited to the case of low speed moving of about 10 km/h as in thepedestrian telephone mode. However, the piloted CDMA can not providehigh quality of communication for a moving unit moving at high speed of100 km/h as in the automobile telephone mode.

Thus, considering the problem in the above-described conventionalsystem, an object of the invention is to provide a large-capacity CDMAtransmission system that can conduct communication with a moving unitsuch as an automobile, transmitting same information quantity as theconventional system without deteriorating communication quality in weakpower transmission in the CDMA and without increasing the occupied bandwidth, using the same frequency band width.

The configuration of a CDMA transmission system for attaining the aboveobject of the invention premises a code division multiple access (CDMA)transmission system in which a primary modulated wave is generated byphase modulation maintaining a phase of a carrier signal at apredetermined phase within a predetermined period of time, spreadsignals are generated by multiplying this primary modulated wave byspread code sequences, and a plurality of spread signals aretransmitted.

According to a first aspect of the invention, the differential codingphase modulation (shift keying) (DPSK) is used for generating theprimary modulated wave on the transmitting side. On the receiving side,quasi-synchronous detection and difference operation are employed todetect the phase difference between the last symbol period and thecurrent symbol period, and the detected phase difference is given as theinformation phase of the current symbol period.

According to a second aspect of the invention, the phases in the ends ofsymbol periods are made to change continuously, in the above-describedcode division multiple access (CDMA) transmission system. By thisoperation, the system is constructed such that a rapid phase change inthe neighborhood area of an end of a symbol period is eliminated.

Further, according to a third aspect of the invention, spread codevalues are changed continuously in an end of a code period (chip) of thementioned spread code sequence in the above-mentioned presumption. Bythis operation, the system is constructed such that a rapid change ofthe spread code values in the neighborhood area of an end of a chipperiod is eliminated.

Further, according to a fourth aspect of the invention, virtual segmentsare superposed in each symbol interval, so that despreading is performedin segments whose number exceeds the number of transmission segments ofeach symbol interval.

Further, various aspects of the present invention may include followingconstruction:

a construction obtained by combining the first aspect with the second,third, or fourth mode;

a construction obtained by combining the first mode with the second,third and fourth aspects;

a construction obtained by combining the first aspect with the secondand third modes, or the second and fourth aspects, or the third andfourth aspects;

a construction obtained by combining the second aspect with the third orfourth aspect;

a construction obtained by combining the second aspect with the thirdand fourth aspects; and

a construction obtained by combining the third aspect with the fourthaspect.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a diagram showing an example of a configuration of adifferential CDMA transmitter according to the present invention;

FIG. 2 is a diagram showing an example of a detailed configuration of adifferential coding circuit in the differential CDMA transmitter shownin FIG. 1;

FIG. 3 is a graph showing an example of characteristics of a π correctorin the differential CDMA transmitter shown in FIG. 1;

FIG. 4 is a diagram showing an example of a configuration of adifferential CDMA receiver according to the present invention;

FIG. 5 is a diagram showing an example of a detailed configuration of adifferential circuit in the differential CDMA receiver shown in FIG. 4;

FIG. 6 is a schematic view showing information phases in the intervalsof symbol 1 and symbol 2 of a primary modulated wave;

FIG. 7 is a diagram showing an example of a detailed configuration of aphase continuous differential coding circuit (DP-CP);

FIG. 8 is a graph showing an example of an operating characteristic of acontinuating circuit (CONTI);

FIG. 9 is a diagram showing an example of a detailed configuration of aphase continuous circuit (CP);

FIG. 10 is a schematic view showing a primary modulated waveform in theintervals of symbol 1 and symbol 2;

FIG. 11 is a schematic view showing a spread code sequence in the chipintervals 1-4;

FIG. 12 is a diagram showing an example of a detailed configuration of aspread code sequence waveform continuating circuit (CODE-CS);

FIG. 13 is graph showing an example of an output characteristic of asmoother (SMO);

FIG. 14 is a view showing an example of an interleaved state of basicsegments and virtual segments;

FIG. 15 is a diagram showing an example of a detailed configuration of avirtual segment interleave despreading circuit (deSS-VSI);

FIG. 16 is a graph showing examples of effect in the stationarytelephone mode of the differential CDMA transmission system according tothe present invention;

FIG. 17 is a graph showing examples of effect in the pedestriantelephone mode of the differential CDMA transmission system according tothe present invention;

FIG. 18 is a graph showing examples of effect in the automobiletelephone mode of the differential CDMA transmission system according tothe present invention;

FIG. 19 is a schematic view showing power spectrum distributions of thepilot channel and the information channel;

FIG. 20 is a graph showing examples of effect in the pedestriantelephone mode in the case that the phase continuous CDMA techniqueaccording to the present invention is applied to the conventional CDMAtransmission system;

FIG. 21 is a graph showing examples of effect in the pedestriantelephone mode in the case that the phase continuous CDMA techniqueaccording to the present invention is applied to the differential CDMAtransmission system;

FIG. 22 is a graph showing examples of effect in the pedestriantelephone mode in the case that the chip waveform continuous CDMAtechnique according to the present invention is applied to thedifferential CDMA transmission system;

FIG. 23 is a graph showing examples of effect in the pedestriantelephone mode in the case that the phase continuous CDMA technique andthe chip waveform continuous CDMA technique according to the presentinvention are applied to the conventional CDMA transmission system;

FIG. 24 is a graph showing examples of effect in the pedestriantelephone mode in the case that the phase continuous CDMA technique andthe chip waveform continuous CDMA technique according to the presentinvention are applied to the differential CDMA transmission system;

FIG. 25 is a graph showing examples of effect in the automobiletelephone mode in the case that the virtual segment interleavedespreading technique according to the present invention is applied tothe differential CDMA transmission system;

FIG. 26 is a diagram showing an example of a configuration of theconventional CDMA transmitter;

FIG. 27 is a schematic view showing a waveform in the intervals ofsymbol 0 and symbol 1 of a primary modulated wave of the CDMAtransmitter shown in FIG. 26;

FIG. 28 shows an example of bit arrangement (bit constellation) for theQPSK;

FIG. 29 shows an example of bit arrangement (bit constellation) for theπ/4-shifted QPSK;

FIG. 30 shows an example of segment configuration in a symbol intervalof a primary modulated wave;

FIG. 31 shows an example of chip configuration in segment intervals;

FIG. 32 is a diagram showing an example of a configuration of theconventional CDMA receiver;

FIG. 33 is a diagram showing an example of a configuration of ademodulator circuit (deMOD) in the CDMA receiver shown in FIG. 32;

FIG. 34 is a diagram showing an example of a configuration of adespreading circuit (deSS) in the CDMA receiver shown in FIG. 32;

FIG. 35 is a diagram showing an example of a configuration of a phasecorrection circuit (CMP) in the CDMA receiver shown in FIG. 32;

FIG. 36 is a graph showing an example of transmission characteristics inthe stationary telephone mode of the conventional CDMA transmissionsystem;

FIG. 37 is a graph showing an example of transmission characteristics inthe pedestrian telephone mode of the conventional CDMA transmissionsystem; and

FIG. 38 is a graph showing an example of transmission characteristics inthe automobile telephone mode of the conventional CDMA transmissionsystem.

BEST MODE FOR CARRYING OUT THE INVENTION

In the following, embodiments of the present invention will be describedreferring to the drawings. In the figures, like numerals or symbolsrefer to like components.

As a first embodiment of the present invention, is proposed a CDMAtransmission system wherein a primary modulated wave obtained by thedifferential coding phase modulation, in which a phase difference in asymbol interval shows information, is spread by a spread code.

As a second embodiment of the present invention, is proposed a phasecontinuous CDMA transmission system wherein a primary modulated waveobtained by phase modulation, in which a phase value continuouslychange, is spread by a spread code sequence.

As a third embodiment of the present invention, is proposed a chipwaveform continuous CDMA transmission system wherein a primary modulatedwave is spread using a chip waveform that is continuously changed.

Further, as a fourth embodiment of the present invention, is proposed avirtual segment interleave CDMA transmission system wherein virtualsegments are set in despreading, and despread code is obtained insegments interleaved and superposed.

In the following, respective features of the present invention will bedescribed with respect to those systems.

[Differential CDMA Transmission System]

As described above, when CDMA communication (in the automobile mode) isconducted moving at a speed of 100 km/h in a city area and using the 2GHz domain, then, the maximum frequency deviation of the Doppler shiftis 200 Hz, the maximum fading frequency is 185 Hz, and propagation delaybetween the first and second propagation paths is 1 μsecond. However,even in a rapid accelerating condition of arriving in 14 seconds at a400 m point from a stationary state, a changing speed of the frequencydeviation due to the Doppler shift is about 30 Hz/sec. and a differencebetween frequency deviations of the Doppler shift in adjacent symbolintervals is as small as 0.001 Hz which can be taken as nearly zero.

Further, although the phase deviation due to fading is as large as ±180degrees, a difference between fading phase deviations in adjacentsymbols is as small as ±0.01 degree which can be taken as nearly zero,also.

Similarly, the propagation delay can be taken as nearly constant betweenadjacent symbol intervals. Namely, a difference between propagationdelays is a quantity decided by a length difference of a propagationpath and a moving speed of the moving unit. Since change of propagationpath length caused-by travelling at 100 km/h in a period of 31.25μseconds of one symbol interval is 0.9 m at maximum, only 3 nanosecondsof difference in propagation delay is generated, and the propagationdelay is nearly constant in adjacent symbol intervals.

It becomes clear that, in the case that information expressed by phasedifferences is transmitted, a phase difference between adjacent symbolsis kept at the value of the time of transmission, even when manyinterfering waves such as reflected waves and diffracted waves are mixedinto the propagation path, and frequency-selective fading is generatedowing to deviation, phase error, and delay error of the carrierfrequency, severely distorting the received wave.

FIG. 1 shows an example of a configuration of a differential CDMAtransmitter according to the present invention, using DQPSK for primarymodulation. The conventional CDMA transmitter of FIG. 26 employs notDQPSK but QPSK for primary modulation. In comparison, the CDMAtransmitter of the present invention has the same configuration as theconventional CDMA transmitter shown in FIG. 26 except that DQPSK isemployed for the primary modulation. In FIG. 1, the information inputterminals 100-10 n, phase modulation circuits (MOD) 110-11 n, spreadingcircuits (SS) 120-12 n, spread code sequence generating circuits (CG)130-13 n, a summing circuit (SUM) 140, a frequency bandlimiting circuit(Band Path Fitter) (BPF) 141, and a transmitting circuit (TX) 142 havethe same functions as the respective components of the same referencenumerals in FIG. 26.

In contrast with the configuration of FIG. 26 in which a primarymodulated wave (QPSK wave) is generated in accordance with inputtedinformation, the feature of the present invention lies in thatdifferential coding circuits (DP) 150-15 n are provided on the inputside of the phase modulation circuits (MOD) 110-11 n, and thosedifferential coding circuits (DP) 150-15 n and the phase modulationcircuits (MOD) 110-11 n constitute a differential coding phasemodulation circuit (diffMOD). This differential coding phase modulationcircuit (diffMOD) generates a primary modulated wave (DQPSK wave)obtained by phase modulation (differential coding phase modulation)using the sum of the phase relating to the input information of thecurrent symbol interval and the phase in the last symbol interval. Inthe following description, DQPSK is used for primary modulation.However, cases in which another differential PSK is used are similar,and description on those cases will be omitted since these cases may beunderstood by analogy.

FIG. 2 shows an example of a detailed configuration of the differentialcoding circuits (DP) 150-15 n placed in the previous stage to the phasemodulation circuits (MOD) 110-11 n. In the figure, an input signal fromthe input terminal 50 is inputted to the adder 51. The adder 51 adds theinput signal and a signal of the feedback from the output 55 of the πcorrector 54. The latch register 52 takes in the output of the adder 51at a leading edge of a clock signal supplied to the clock terminal (CLK)53, to hold and then input it to the π corrector 54.

Here, the π corrector (NCOR) 54 has the input/output characteristicsshown in FIG. 3, and, when an input value a is −π or more or π or less,outputs the value a. When an input value a exceeds π, a value a−2π isoutputted, and when an input value a is less than −π, a value a+2π isoutputted.

Returning to FIG. 2, operation of the differential coding circuits (DP)150-15 n will be described. It is assumed that the input 50 holds phaseinformation a of the next symbol and the latch register 52 holds adifferential coding phase value b of the current symbol. However, forthe sake of convenience, it is assumed that the absolute value of b isless than π.

Although the output of the adder 51 is a+b, the value b is held until aclock is applied to the terminal 53. After it is held for a periodcorresponding to the symbol interval T, when a clock is applied to theclock terminal 53, the latch register 52 takes in the value a+b at theleading edge of the clock signal, and holds the value until the nextclock is applied. Further, at the same time with the clock, the input 50is updated, to change to a phase value c of the next symbol.

In the meantime, the π corrector 54 judges the value of a+b and selectsa value having the minimum absolute value out of three values a+b, anda+b±2π, to output it. Since a trigonometric function sin or cos providesthe same function value for any of the three values, a+b−2π, a+b,a+b+2π, the primary modulated wave after the phase modulation shows thesame waveform. By utilizing this property such that values stored orprocessed in the differential coding circuits (DP) 150-15 n becomewithin an interval between −π and +π, the processing circuits can beprevented from becoming complex.

FIG. 4 shows an example of a configuration of a receiver to which thedifferential CDMA according to the present invention is applied. In thefigure, the receiving circuit (RX) 200, demodulator circuit (deMOD) 201,input terminal 202 for a demodulation control signal, synchronismdetection circuit (SYNC) 203 , reception control detection circuit (CNT)204 , despreading circuits (deSS) 210-21 n, input terminals 220-22 n fordespread code sequences, decision circuits (DEC) 240-24 n, and outputterminals 250-25 n have the same functions as the respective componentsadded with the corresponding reference numerals of the receiver of FIG.32 to which the conventional CDMA is applied, and their detaileddescription will be omitted.

Differently from the conventional configuration of FIG. 32, thedifferential circuits (DIFF) 260-26 n are substitute for the phasecorrection circuits (CMP) between the despreading circuits (deSS) 210-21n and the decision circuits (DEC) 240-24 n.

FIG. 5 shows an example of a configuration of such differential circuits(DIF) 260-26 n. In the figure, the in-phase component I_(i)′(t) andquadrature component Q_(i)′(t) of a despread signal as an output of thedespreading circuit (deSS) are inputted to the input terminals 2500,2501. I_(i)′(t) is inputted to the multipliers 2503 and 2507, and to thedelay circuit 2502. Q_(i)′(t) is inputted to the multipliers 2504 and2506, and to the delay circuit 2505. To the multipliers 2503 and 2504,is inputted the in-phase component I_(i)′(t) of the despread signalthrough the delay circuit 2502. To the multipliers 2506 and 2507, isinputted the quadrature component Q_(i)′(t) of the despread signalthrough the delay circuit 2505. Here, the delay circuits 2502, 2505delay an input f(t) only by a period T corresponding to one symbolinterval to output a signal f(t−T).

The adder 2508 adds the outputs of the multipliers 2503 and 2506, andoutputs the result as the in-phase component I_(i)(t) of thedifferential signal to the output terminal 2510. Further, the adder 2509subtracts the output of the multiplier 2507 from the output of themultiplier 2504 to output the result as the quadrature componentQ_(i)(t) of the differential signal to the output terminal 2511.

In the configuration of the embodiment of FIG. 4, the differentialcircuits (DIFF) 260-26 n are placed after the demodulator circuit(deMOD) 201 and the despreading circuits (deSS) 210-21 n. However, everyprocessing in a series of these circuits is a linear operation, and theorder of processing does not affects the actions. Thus, it can be easilyinferred that the order of placing the differential circuits (DIFF)260-26 n, demodulator circuit (deMOD) 201 , and the despreading circuits(deSS) 210-21 n does not affect the processing result. Thus, locationsof the various circuits are not limited to the configuration of FIG. 4.

The output i(t) of the in-phase component and the output q(t) of thequadrature component of the demodulator circuit (deMOD) 201 include aplurality of received wave components arriving through multi-raypropagation path as described above in relation to FIG. 32. Accordingly,it is inevitably impossible to completely synchronize a plurality ofcarrier waves included in the received wave and a regenerated carrierwave regenerated by the synchronism detection circuit (SYNC) 203 . Thus,synchronous detection in the demodulation processing becomes incomplete,and the conventional CDMA receiver includes many errors, which is acause of the poor communication characteristics. However, in theconventional CDMA reception, such incomplete synchronous detection istaken as the complete synchronous detection without including an error,which is a cause of deteriorating communication quality. On the otherhand, the present invention is designed to exclude such errors owing toincomplete synchronization. By this reason, in the techniques disclosedin the present invention, synchronous detection including an error inthe carrier frequency, phase, or delay is called quasi-synchronousdetection differentiating it from the complete synchronous detection, tomake the existence of error clear.

In the differential CDMA reception, the demodulated signal by thequasi-synchronous detection is despread to obtain in-phase componentsand quadrature components of (n+1) despread signals. With respect to aseries of processing for obtaining the (n+1) despread signals from thereceived wave, the conventional CDMA reception and the differential CDMAreception disclosed here are same to each other.

However, in the conventional CDMA reception, not all the (n+1) despreadsignals are used for transmitting information, but at least one is usedas a common pilot signal and only n remaining despread signals are usedfor transmitting the information. Namely, a known value is transmittedas a pilot. A phase error of the pilot appearing at the time ofreception is taken as a disturbance suffered in the course ofpropagation. The (n+1) channels are assumed to suffer the same phaseerror, and, thus, phase correction is carried out commonly on theremaining n despread signals, in order to receive n pieces ofinformation.

On the other hand, the differential CDMA reception disclosed here isdesigned such that all the (n+1) despread signals are used fortransmitting information, that, by obtaining a difference betweenreceived phases of adjacent symbols in each channel, the effect oferrors of the quasi-synchronous detection is excluded, and theinformation can be received without being affected by thefrequency-selective fading. Namely, the differential circuit (DIFF) 26 iof the channel i carries out the difference operation shown in thefollowing, using the despread signal of the channel i, the in-phasecomponent I_(i)′(t) and quadrature component Q_(i)′(t) of the despreadsignal, and the in-phase component I_(i)′(t−T) and quadrature componentQ_(i)′(t−T) of the despread signal delayed by one symbol period T, andoutputs the results as the in-phase component Î_(i)(t) and quadraturecomponent {circumflex over (Q)}_(i)(t) of the phase differential signal.

The in-phase component Î_(i)(t) and quadrature component {circumflexover (Q)}_(i)(t) of the phase differential signal, shown in theequations 35 and 36,Î _(i)(t)=I _(i)′(t)I _(i)′(t−T)+Q _(i)′(t)Q _(i)′(t−T)  (35){circumflex over (Q)} _(i)(t)=−I _(i)′(t)Q _(i)′(t−T)+Q _(i)′(t)I_(i)′(t−T)  (36)can be obtained by substituting the in-phase component I_(i)′(t) andquadrature component Q_(i)′(t) of the despread signal, given by theequations 19 and 20, and the delayed in-phase component I_(i)′(t−T) anddelayed quadrature component Q_(i)′(t−T) into those equations 35 and 36.In order to make the comparison clear, these components are obtained bysubstituting the in-phase component I_(i)′(t) and quadrature componentQ_(i)′(t) of the despread signal under the two-ray Rayleigh fadingenvironment, given by the equations 32 and 33, and the delayed in-phasecomponent I_(i)′(t−T) and the delayed quadrature component Q_(i)′(t−T),similarly to the description of the conventional technique, as follows.$\begin{matrix}\begin{matrix}{{{\hat{I}}_{i}\quad(t)} = {\overset{\sim}{\beta}\quad(t)\quad\overset{\sim}{\beta}\quad\left( t^{\prime} \right)\quad\cos\left\{ {{\theta_{i}\quad\left( {t - {\delta_{1}\quad(t)}} \right)} + {{\overset{\sim}{\varphi}}_{i}\quad(t)} +} \right.}} \\{{\left. {\overset{\sim}{\vartheta}\quad(t)} \right\}\quad\cos\left\{ {{\theta_{i}\quad\left( {t^{\prime} - {\delta_{1}\quad\left( t^{\prime} \right)}} \right)} + {{\overset{\sim}{\varphi}}_{i}\quad\left( t^{\prime} \right)} + {\overset{\sim}{\vartheta}\quad\left( t^{\prime} \right)}} \right\}} +} \\{\overset{\sim}{\beta}\quad(t)\quad\overset{\sim}{\beta}\quad\left( t^{\prime} \right)\quad\sin\left\{ {{\theta_{i}\quad\left( {t - {\delta_{i}\quad(t)}} \right)} + {{\overset{\sim}{\varphi}}_{i}\quad(t)} + {\overset{\sim}{\vartheta}\quad(t)}} \right\}\quad\sin\left\{ {{\theta_{i}\quad\left( {t^{\prime} - {\delta_{i}\quad\left( t^{\prime} \right)}} \right)} +} \right.} \\\left. {{{\overset{\sim}{\varphi}}_{i}\quad\left( t^{\prime} \right)} + {\overset{\sim}{\vartheta}\quad\left( t^{\prime} \right)}} \right\} \\{= {\overset{\sim}{\beta}\quad(t)\quad\overset{\sim}{\beta}\quad\left( t^{\prime} \right)\quad\cos\left\{ {{\theta_{i}\quad\left( {t - {\delta_{i}\quad(t)}} \right)} - {\theta_{i}\quad\left( {t^{\prime} - {\delta_{i}\quad\left( t^{\prime} \right)}} \right)} + {{\overset{\sim}{\varphi}}_{i}\quad(t)} - {{\overset{\sim}{\varphi}}_{i}\quad\left( t^{\prime} \right)} +} \right.}} \\\left. {{\overset{\sim}{\vartheta}\quad(t)} - {\overset{\sim}{\vartheta}\quad\left( t^{\prime} \right)}} \right\}\end{matrix} & (37)\end{matrix}$ $\begin{matrix}\begin{matrix}{{{\hat{Q}}_{i}\quad(t)} = {\overset{\sim}{\beta}\quad(t)\quad\overset{\sim}{\beta}\quad\left( t^{\prime} \right)\quad\cos\left\{ {{\theta_{i}\quad\left( {t - {\delta_{1}\quad(t)}} \right)} + {{\overset{\sim}{\varphi}}_{i}\quad(t)} +} \right.}} \\{{\left. {\overset{\sim}{\vartheta}\quad(t)} \right\}\quad\sin\left\{ {{\theta_{i}\quad\left( {t^{\prime} - {\delta_{1}\quad\left( t^{\prime} \right)}} \right)} + {{\overset{\sim}{\varphi}}_{i}\quad\left( t^{\prime} \right)} + {\overset{\sim}{\vartheta}\quad\left( t^{\prime} \right)}} \right\}} +} \\{\overset{\sim}{\beta}\quad(t)\quad\overset{\sim}{\beta}\quad\left( t^{\prime} \right)\quad\sin\left\{ {{\theta_{i}\quad\left( {t - {\delta_{i}\quad(t)}} \right)} + {{\overset{\sim}{\varphi}}_{i}\quad(t)} + {\overset{\sim}{\vartheta}\quad(t)}} \right\}\quad\cos\left\{ {{\theta_{i}\quad\left( {t^{\prime} - {\delta_{i}\quad(t)}} \right)} +} \right.} \\\left. {{{\overset{\sim}{\varphi}}_{i}\quad\left( t^{\prime} \right)} + {\overset{\sim}{\vartheta}\quad\left( t^{\prime} \right)}} \right\} \\{= {\overset{\sim}{\beta}\quad(t)\quad\overset{\sim}{\beta}\quad\left( t^{\prime} \right)\quad\sin\left\{ {{\theta_{i}\quad\left( {t - {\delta_{i}\quad(t)}} \right)} - {\theta_{i}\quad\left( {t^{\prime} - {\delta_{i}\quad\left( t^{\prime} \right)}} \right)} + {{\overset{\sim}{\varphi}}_{i}\quad(t)} - {{\overset{\sim}{\varphi}}_{i}\quad\left( t^{\prime} \right)} +} \right.}} \\\left. {{\overset{\sim}{\vartheta}\quad(t)} - {\overset{\sim}{\vartheta}\quad\left( t^{\prime} \right)}} \right\}\end{matrix} & (38)\end{matrix}$where t′=t−T.

In adjacent symbol intervals, as already described, fading, Dopplershift, and the like are nearly constant, and thus, the followingequations can be obtained.Î _(i)(t)≅{tilde over (β)}(t){tilde over (β)}(t−T)cos {θ_(i)(t−δ₁(t))−θ₁(t−T−δ ₁(t−T))}  (39){circumflex over (Q)} _(i)(t)≅{tilde over (β)}(t){tilde over(β)}(t−T)sin {θ_(i)(t−δ ₁(t))−θ_(i)(t−T−δ ₁(t−T))}  (40)

As already described in relation to the conventional technique, in manycases, the decision circuit obtains the received information from thereceived phase angle. In the conventional technique, the phase anglebetween the in-phase component and the quadrature component of the phasecorrection signal is obtained. On the other hand, as a feature of thepresent invention, the phase angle between the in-phase component andquadrature component of the phase differential signal is obtained.Namely,$\begin{matrix}\begin{matrix}{{{information}\quad{phase}_{i}\quad(t)} = {\tan^{- 1}\left\lbrack \frac{{\hat{Q}}_{i}\quad(t)}{{\hat{I}}_{i}\quad(t)} \right\rbrack}} \\{= {\tan^{- 1}\left\lbrack \frac{\sin\left\{ {{\theta_{i}\quad\left( {t - {\delta_{q}\quad(t)}} \right)} - {\theta_{i}\quad\left( {t - T - {\delta_{1}\quad\left( {t - T} \right)}} \right)}} \right\}}{\cos\left\{ {{\theta_{i}\quad\left( {t - {\delta_{1}\quad(t)}} \right)} - {\theta_{i}\quad\left( {t - T - {\delta_{1}\quad\left( {t - T} \right)}} \right)}} \right\}} \right\rbrack}} \\{= {{\theta_{i}\quad\left( {t - {\delta_{i}\quad(t)}} \right)} - {\theta_{i}\quad\left( {t - T - {\delta_{1}\quad\left( {t - T} \right)}} \right)}}}\end{matrix} & (41)\end{matrix}$

It can be seen that the last right side of the equation 41 gives thedifference between the phase angle θ_(i)(t−δ(t)) in the current symbolinterval and the phase angle θ_(i)(t−T−δ(t−T)) in the previous symbolinterval. In the present invention, information is changed to adifferential code before transmission, and thus it becomes obvious thatthe phase difference θ_(i)(t−δ(t))−θ_(i)(t−T−δ(t−T)) correctly receivesthe received information of the channel i. In the case of the last rightside of the equation 34 expressing the information phase angle accordingto the conventional system, the Doppler shift, fading phase error,carrier regeneration delay, and the like are removed. However, thefrequency-selective fading distortion{tilde over (ψ)}_(i)(t)−{tilde over (ψ)}₀(t)can not be removed, remaining as a disturbance term. Due to thisdisturbance term, a reception error arises, deteriorating thecommunication quality. On the other hand, in the present invention, itis obvious that, as shown in the equation 41, only the phase difference,i.e. the information itself, remains in the information phase angleobtained from the phase differential signal, and a disturbance termother than the information is completely removed.

Further, when the information phase is obtained using the averages inthe symbols in the in-phase component and quadrature component of thephase differential signal shown in the equations 39 and 40, randomnoises can be suppressed and the communication quality can be improved.This can be easily understood, and therefore its description is omitted.

Further, the envelope of the phase differential signal can be obtainedby the square of the square-sum of the in-phase component and quadraturecomponent of the phase differential signal. For detecting the amplitudedistortion etc., it is sufficient to use this envelope of the phasedifferential signal. Further, the amplitude distortion can be easilyremoved by normalizing the phase differential signal with the envelope.This can be easily understood, and therefore its description is omitted.

[Phase Continuous CDMA Transmission System]

It is known that the frequency band of the received wave, which ispropagated being affected by the fading in the course of thetransmission path, becomes larger than the original frequency band ofthe radio wave emitted from a transmitter. This enlarged frequency bandwidth is called a fading band width.

As shown in equations 19 and 20, in the variables of the trigonometricfunctions expressing the in-phase component and quadrature component ofthe despread signal, other than the information θ_(i), the term {tildeover (ψ)}_(i)(t) relating to the frequency-selective fading and the termrelating to the fading phase error, Doppler shift, carrier regenerationerror, and the like become factors of enlarging the frequency bandwidth.

The higher the speed of a moving unit becomes, the higher the fadingfrequency becomes. And, to that extent, the fading band width isenlarged, and as a result, communication quality is deteriorated. As themoving speed of a moving unit increases from the stationary telephonemode through the pedestrian telephone mode to the automobile telephonemode, the communication quality is deteriorated to that extent. In fact,as shown in FIG. 36, in the stationary telephone mode, the receivedelectric field required for obtaining BER≦0.001 can be realized for allthe transmission band width of 1.51 MHz or more at Eb/No≦0 dB.

On the other hand, as shown in FIG. 37, in the pedestrian telephonemode, the transmission band width required for obtaining BER≦0.001 is3.46 MHz or more for the same received electric field of Eb/No≦0 dB, anda floor appears for the transmission band width of 3.37 MHz. In thefollowing, the minimum transmission band width realizing the highquality communication of this BER≦0.001 is called the criticaltransmission band width, which is used for evaluating a CDMAtransmission system.

As the critical transmission band width is smaller, it means that thefrequency utilization efficiency is better. Effective utilization of thelimited frequency resource is considered to be an important factor forevaluating a system, and thus, the critical transmission band width isused for evaluating a system.

As shown in FIG. 37, in the automobile telephone mode, BER is always 0.2or more, and accordingly, the communication quality can not be improvedeven by the highest transmission power. In this case, the criticaltransmission band width is more than 25.60 MHz. As the cause of suchoccurrence of a floor, it is mentioned that the frequency band width ofthe transmission wave exceeds the allowable transmission band width, dueto the increase of the fading band width.

The transmission band width is determined in advance, at the time of thesystem design, while the fading band width is determined by the movingspeed of a moving unit, etc. Thus, for realizing high qualitycommunication in the automobile telephone mode, it is efficient means tomake the band width of the transmission wave narrower, to have a largermargin for increase of the fading band width.

The band width of the transmission wave is defined by convolution of thefrequency band width of a spread code sequence and the frequency bandwidth of a primary modulated wave. When information is not carried, aprimary modulated wave becomes a tone signal of the carrier frequency,and the band width becomes zero, so that the frequency band width of thetransmission wave coincides with the band width of the spread codesequence.

On the other hand, when information is carried, a primary modulated wavehas discontinuity of phase at ends of a symbol interval. Thisdiscontinuity of phase enlarges the frequency band width, andconvolution is also increased, so that the frequency band width of thetransmission wave becomes wideband. In that case, information istransmitted by a phase value within a symbol period of the primarymodulated wave, and not by violent phase fluctuation such as phasediscontinuity at ends of a symbol interval. Accordingly, when violentphase change at the ends of a symbol interval is excluded, high qualitycommunication can be realized without enlarging the transmission bandwidth.

FIG. 6 shows an example of phases in the symbol interval 1 and symbolinterval 2 is shown by a solid line. In the figure, phases relating toinformation of the conventional primary modulated wave are shown by abroken line. Further, in the symbols of the figure, is shown an examplein which the symbol 1 is −π/4 radian and the symbol 2 is π/4 radian.

As shown by the broken line, when an information phase is defined allover the symbol interval, extreme change (discontinuity) of the primarymodulated wave appears at an end of the symbol interval as shown in FIG.27, which becomes a cause of increasing the frequency band width of theprimary modulated wave.

In contrast, according to the phase continuous technique as a feature ofthe present invention, a transition interval is provided in theneighborhood of each end of a symbol interval as shown by the solid lineof FIG. 6, and the information phase changes continuously in thistransition interval. Here, as shown by a solid curve in FIG. 6, thetransition interval 1 refers to an interval set from the time−ΔT/2 tothe time ΔT/2 between the symbol 0 and symbol 1, in which the phasechanges continuously; the transition interval 2 to an interval set fromthe time T−ΔT/2 to the time T+ΔT/2 between the symbol 1 and the symbol2, in which the phase changes continuously; and the transition intervalk+1 to an interval set from the time kT−ΔT/2 to the time kT+ΔT/2 betweenthe symbol k and the symbol k+1. Further, the length of each transitioninterval is a constant ΔT.

FIG. 7 is a block diagram showing an example of configuration of thephase continuous differential coding circuit (DP-CP) used, in place ofthe differential coding circuit (DP) 150-15 n of FIG. 1, in thedifferential CDMA transmitter according to the present invention. In thefigure, a signal inputted from the input terminal 500 is inputted to theadder 501. The adder 501 adds the input signal from the input terminal500 and a shift signal from the shift-constant generating circuit(OFFSET) 502. The output of the adder 501 is inputted to the π corrector(πCOR) 503. The output 504 of the π corrector (πCOR) 503 and the outputof the continuating circuit (CONTI) 506 are inputted to the multiplier505, which outputs the product of both the inputs. Further, the output511 of the phase continuous differential coding circuit (DP-CP) isreturned in feedback through the 2π corrector (2πCOR) 510 and the latchregister (REG) 508, and added with the output of the multiplier 505 bythe adder 507, to become the output 511. Further, in FIG. 7, at aleading edge of the clock signal inputted to the terminal (CLK) 509, thelatch register (REG) 508 takes in its input to hold therein.

Here, the 2π corrector (2πCOR) has terminals for input a and output b.When a value inputted to the terminal a exceeds 2π(a>2π), the 2πcorrector outputs a value (a−2π) to the output terminal b. When a valueinputted to the terminal a is less than −2π(a<−2π), it outputs a value(a+2π). And, when a value inputted to the terminal a is more than orequal to −2π and less than or equal to 2π(−2π≦a≦2π), it outputs theinput value a itself. Further, as shown in FIG. 8, the continuatingcircuit (CONTI) outputs a value that changes continuously from 0 to 1 inthe transition interval k, for example, the value defined by thefollowing equation 42. $\begin{matrix}{{\frac{1}{2}\left\{ {1 + {\sin\quad\left( {\pi\quad\frac{t - {kT}}{\Delta\quad T}} \right)}} \right\}},{{{t - {kT}} \leq {\Delta\quad T}}}} & (42)\end{matrix}$Since an transition interval exists for each period of time T, this canbe easily realized, for example, by storing in a ROM in advance theoutput values of the continuating circuit, and by reading these valuesin turn from the ROM, such that a circuit of these values can be made inone symbol interval.

Returning to FIGS. 6 and 7, the operation of the phase continuousdifferential coding circuit (DP-CP) will be described in due order.Since the operation is cyclic in the interval T, the operation will bedescribed from the trailing edge t=ΔT/2 of the transition interval 0 tothe trailing edge t=T+ΔT/2 of the transition interval 1. However, theoperation is similar in the other periods. It is assumed that, at thetime t=ΔT/2, a phase value a₂ of the symbol 2 is applied to the inputterminal 500, and a differential coding phase value b₁ of the symbol 1is outputted to the output terminal 511. However, for the sake ofconvenience, it is assumed that the absolute value of b₁ is less than orequal to 2π.

At the time t=ΔT/2, the latch register 508 latches therein the output b₁of the 2π corrector 510. The latched value is held therein until thetime t=T+ΔT/2 when a clock is inputted to the terminal 509 next time. Inthe case of shiftDPSK, the corresponding shift quantity is stored intothe shift-constant generating circuit (OFFSET) 502. In the case of DPSK,the value “0” is stored in the shift-constant generating circuit 502. Inthe following, the value stored in the shift-constant generating circuit(OFFSET) 502 is uniformly written as d.

Thus, the output d of the shift-constant generating circuit (OFFSET) 502is added to the input a₂ from the input terminal 500, and the result isinputted to the π corrector (πCOR) 503. The π corrector (πCOR) 503judges the output a₂+d of the adder 501, to select a value having theminimum absolute value out of three values a₂+d, a₂+d−2π, and a₂+d+2π,and outputs the selected value to the output terminal 504. This outputis referred to as p_(add).

The multiplier 505 outputs the product of p_(add) and the output of thecontinuating circuit (CONTI) 506. The sum of the output of themultiplier 505 and the value b₁ held by the latch register 508 appearsat the output terminal 511. In the transition interval 2, the output ofthe continuating circuit continuously changes from the value b₁ toarrive at a value b₁+p_(add) at the time t=T+ΔT/2. At that time, whilethe output of 2π corrector (2πCOR) 510 is changing in the transitioninterval 2, the output is finally decided as a value having the minimumabsolute value out of three values b₁+p_(add), b₁+p_(add)−2π,b₁+p_(add)+2π at the trailing edge t=T+ΔT/2. This value will be writtenas b₂.

Further, at the time t=T+ΔT/2, when the next clock is applied, the latchregister 508 takes in the value b₂ to hold therein, and the phase valuea₃ of the symbol 3 is applied to the input terminal 500 and heldthereat.

As described above, to the output terminal 511 of the phase continuousdifferential coding circuit (DP-CP), is outputted the phase signal thathas been subjected to differential coding and changes continuously.Accordingly, by substituting the phase continuous differential codingcircuit (DP-CP) shown in FIG. 7 for each of the differential codingcircuits (DP) 150-15n, the phase continuous technique in thedifferential CDMA transmission system according to the present inventionis realized.

FIG. 9 is a block diagram showing an example of a configuration of thephase continuous circuit in the case that the phase continuous techniqueaccording to the present invention is applied to the conventional CDMAtransmission system shown in FIG. 18. Namely, by placing this phasecontinuous circuit (CP) in the previous stage to each of the phasemodulation circuits (MOD) 110-11 n in FIG. 26, is realized a newtransmission system in which the phase in the transition interval of aprimary modulated wave by the conventional CDMA transmission system canchange continuously. In FIG. 9, the shift-constant generating circuit(OFFSET) 523, the continuous circuit (CONTI) 526, and the latch register(REG) 528 have the same functions as the shift-constant generatingcircuit (OFFSET) 502, the continuous circuit (CONTI) 506, and the latchregister (REG) 508 of FIG. 7, respectively. At the leading edge of theclock signal applied to the clock terminal (CLK) 530, the latch register(REG) 528 takes in an input, and hold therein. The operation of thephase continuous circuit will be described in due order, referring toFIGS. 6 and 9. The operation will be described from the last edge t=ΔT/2of the transition interval 0 to the transition interval t=T+ΔT/2.However, the operation is cyclic in the transition interval. Thus, sincethe operation in the other intervals is similar and can be easilyunderstood, description of such operation will be omitted. It is assumedthat, at the time t=ΔT/2, the phase value a₂ of the symbol 2 is inputtedto the input terminal 520, and the phase value a₁ of the symbol 1 isoutputted to the output terminal 529. At the leading edge of a clockapplied at the time t=ΔT/2, the latch register (REG) 528 latches thevalue a₁ inputted to the input terminal and holds that value until thenext clock is applied at the time t=T+ΔT/2. The sum of the input a₂ andthe output d of the shift-constant generating circuit (OFFSET) 502 isoutputted from the adder 521. Next, the adder 524 outputs the differencebetween that sum and a₁ held by the latch register (REG) 528. Thisdifference corresponds to the phase difference between the symbol 1 andthe symbol 2.

As shown in FIG. 8, the output of the continuating circuit (CONTI) 526changes continuously from 0 to 1 in a transition interval, similarly tothe description given to the continuating circuit (CONTI) 506 of FIG. 7.Namely, as the product of the phase difference and the output of thecontinuating circuit (CONTI) 526, the multiplier 525 outputs the value 0at the leading edge t=T−ΔT/2 of the transition interval 1 and the valuea₂−a₁ at the last edge t=T+ΔT/2 of the transition interval 1. In thisperiod, the output of the multiplier 525 continuously changes from 0 toa₂−a₁. The adder 527 obtains the sum of the output by the multiplier 525and the value a₁ held in the latch register (REG) 528, and outputs theresult to the output terminal 529, so that the output of thecontinuating circuit (CONTI) in the transition interval 2 continuouslychanges from a₁ to a₂. Further, at the last edge t=T+ΔT/2 of thetransition interval 1, the output of the output terminal 529 is taken inby the latch register (REG) 528. Since the adder 527 outputs to theoutput terminal 529, the value a₂ held by the latch register (REG) 528is outputted from the output terminal without interruption even when theoutput of the multiplier 525 becomes 0 at the last edge of thetransition interval 2.

Thus, by placing the phase continuous circuit shown in FIG. 9 previouslyto the phase modulation circuit (MOD) in FIG. 26 from a conventionalCDMA transmission system, is realized a new transmission system in whichthe phase changes continuously in a transition interval of a primarymodulated wave.

As described above, by making the information phase change continuously,discontinuity of a primary modulated wave can be eliminated as shown bythe solid line in FIG. 10, which can suppress increase of the band widthof the modulated wave. For comparison, the broken line in FIG. 10 showsthe conventional primary modulated wave in which phases are notcontinuous. The smooth continuity of the primary modulated wave shown bythe solid line in FIG. 10 can be realized since the discontinuity shownin FIG. 6 is excluded according to the present invention.

[Chip Waveform Continuous CDMA Transmission System]

In the conventional CDMA transmission system or the differential CDMAtransmission system disclosed by the present invention, a PSK or DPSKwave of a primary modulated wave is multiplied by a spread code sequencesuch as a Walsh code sequence, to generate a spread spectrum signal. Inthe schematic view of FIG. 11 showing a time response waveform of aspread code sequence in the chips 1-4, a broken line shows an example ofthe conventional waveform of a spread code sequence. Although FIG. 11shows a case in which both intervals of the chip 1 and chip 2 have acode value 1, the interval of chip 3 has a chip value −1, and theinterval of chip 4 has a code value 1, the other cases are similar. Inthe case that values of spread codes are same between adjacent chipintervals such as the chip intervals 1 and 2, discontinuity in waveformdoes not arise between the adjacent chip intervals.

On the other hand, in the case that values of spread codes are differentfrom each other as between the chip interval 2 and chip interval 3, orbetween the chip interval 3 and chip interval 4, drastic change ofwaveform of the spread signal appears at an end of a chip interval.

When a code value is maintained all over the chip interval to the end ofthe chip interval, drastic change of the waveform occurs at ends of chipintervals as shown in the broken lines of FIG. 11, and the frequencyband width of the spread signal increases extremely. On the contrary,when the drastic fluctuation of the waveform of the spread code sequenceat ends of chip intervals is excluded, increase of the frequency bandwidth of the spread signal can be prevented. When the smoothness of thespread code sequence waveform is made maximum, the maximum frequencyband width appears in the case of alternate pattern of code values, andits frequency band width is given by a reciprocal 1/2τ of the length 2τof two chip intervals. The frequency band width of a spread codesequence decreases in inverse proportion to probability that code valuesare same between adjacent chip intervals, arriving at minimum value 0when chip values are same in all the chip intervals.

However, in order to maintain orthogonality of spread code sequences, itis necessary to maintain the value of a spread code as long as possiblewithin a chip interval, and the frequency band width of 1/2τ isinsufficient for that purpose. On the other hand, when a code value ismaintained all over the chip interval as shown by the broken waveform ofFIG. 11, the frequency band width increases more than necessarily. Inthe case that a radio wave propagates through poor transmission pathfilled up with noise as in mobile communication, even when transmissionis carried out while a code value is maintained all over the chipinterval, received chip waveform gets largely out of shape due to randomnoise entering the transmission path. Further, in order to conductcommunication in a predetermined band width, it is necessary to limitthe frequency band width of the transmission wave. Owing to the bandwidth restriction; sharp waveforms at ends of chip intervals areinevitably lost. Further, as a more important problem, the effect of theband width restriction does not only distort the waveform of the spreadcode sequence, but also extends to the carrier waveform. As a result,even the carrier waveform is distorted and information phases to bepropagated change, which is a cause of deterioration of communicationquality.

Considering such background, suppressing unnecessary sharp fluctuationof the spread code sequence waveform is an important problem.Accordingly, the present invention is designed such that, as shown bythe solid lines of FIG. 11, only when spread code values are differentbetween adjacent chip intervals, slow change is made to occur in atransient interval. Here, as shown in FIG. 11, the transient intervalmeans an interval provided in the neighborhood of each end of chipinterval to extend over adjacent chip intervals. For the sake ofconvenience, the transient interval between the chip intervals 0 and 1is called the transient interval 1, the transient interval between thechip intervals 1 and 2 is called the transient interval 2, and so on.Further, it is assumed that all the transient intervals have the sametime length R.

FIG. 12 is a block diagram showing an example of a configuration of acircuit that realizes chip waveform continuating. This spread codewaveform continuating circuit (CODE-CS) is inserted, for example,between each pair of the spread code generating circuits (CG) 130-13 mand the spreading circuits (SS) 120-12 n of FIG. 1.

In FIG. 12, a spread code sequence from the corresponding spread codegenerating circuit (CG) 13 i is inputted to the input terminal 300, andas itself inputted to the adder 301. At a leading edge of the clocksignal applied to the clock terminal (CLK) 307, the output from theoutput terminal 306 of the spread code sequence waveform continuatingcircuit (CODE-CS) is taken in to the latch register (REG) 305 to be heldtherein. The adder 301 outputs a difference between the value of thespread code inputted to the input terminal 300 and the value held in thelatch register (REG) 305. The multiplier 302 outputs the product of theoutput of the adder 301 and a value outputted from the smoother (SMO)303. Next, the output of the adder 302 is added to the output of thelatch register 305 in the adder 304, and the obtained sum is outputtedto the output terminal 306 of the spread code sequence waveformcontinuating circuit (CODE-CS).

FIG. 13 is a graph showing a time response of the output of the smoother(SMO) 303, and the time response, smoother(t) continuously changes from0 to 1 in each transient interval, as shown by the following equation.$\begin{matrix}{{{{smoother}\quad(t)} = {\frac{1}{2}\left\{ {1 + {\sin\quad\left( {\pi\quad\frac{t - {k\quad\tau}}{R}} \right)}} \right\}}},{{{t - {k\quad\tau}}} \leq R}} & (43)\end{matrix}$

As shown in the equation 43, the output of the smoother gives valuescyclic in the chip interval T, and thus, the smoother can be realized bycyclicly reading a ROM or the like storing the values for one chipinterval.

Returning to FIGS. 11 and 12, the operation of the spread code waveformcontinuating circuit (CODE-CS) will be described in due order. Theoperation is cyclic in the chip interval, description is given withrespect to the period from the time t=R/2 to the time t=2τ+R/2. Theoperations in the other chip intervals are similar and can be easilyunderstood, and their description is omitted. It is assumed that, at thetrailing edge of the transient interval t=R/2, the clock is applied tothe clock terminal (CLK) 307, and the spread code value in the next chipinterval is decided. As shown in FIG. 11, the output of the outputterminal 306 is 1 at the time t=R/2, and thus, the latch register (REG)305 takes in 1 to hold therein. In addition, the spread code value 1 ofthe chip interval 2 is applied to the input terminal 300. Since theoutput of the latch register (REG) 305 and the input of the inputterminal 300 are same, the output of the adder 301 becomes 0.Accordingly, regardless of the output value of the smoother (SMO) 303,the output of the multiplier 302 is 0 all over the transient interval 2,the output of the adder 304 does not change from 1, and the value 1 iscontinuously outputted to the output terminal 305. As shown in FIG. 11,the spread code sequence waveform retains the value 1. Further, at thetrailing edge t=τ+R/2 of the transient interval 2, the latch register(REG) 305 takes in the output value 1 to hold therein, and the spreadcode value −1 of the next chip interval is applied to the inputterminal.

Accordingly, although the output of the adder 301 becomes −2, the outputof the smoother (SMO) 303 is 0 until the leading edge of the transientinterval 3, and the multiplier 302 continuously outputs the value 0.However, since the sum of the multiplier 304 and the latch register(REG) 305 is outputted to the output terminal 306, the value 1 held inthe latch register 305 is continuously outputted until the leading edgeof the transient interval 3. In the transient interval 3, at the leadingedge, the output of the smoother 303 rises from the value 0, andcontinuously increases to the value 1 at the last edge. Accordingly, theoutput of the multiplier 302 changes in the range from 0 to −2. By this,the sum of the output of the multiplier 304 and the value held in thelatch register 305 appears at the output terminal, changing in the range1-(−1). Accordingly, the spread code value in the transient interval 3is shaped into a smoothly changing waveform as shown in FIG. 11.Further, at the last edge of the transient interval 3, the value −1 ofthe output terminal is taken in by the latch register 305, and theoperation moves to the next stage, in which the operation similar to theabove is carried out.

As described above, by using the spread code sequence waveformcontinuating circuit (CODE-CS), the chip waveform of the spread signalis made continuous. By inserting this spread code sequence waveformcontinuating circuit (CODE-CS) between each pair of the spread codegenerating circuits (CG) 130-13 n and the spreading circuits (SS) 120-12n of FIG. 1, is realized the differential CDMA transmission system towhich the chip waveform continuating technique of the present inventionis applied. Or, in FIG. 29 showing the conventional CDMA transmissionsystem, by inserting the spread code waveform continuating circuit(CODE-CS) between each pair of the spread code generating circuits130-13 n and the spreading circuits 120-12 n, can be realized the CDMAtransmission system to which the chip waveform continuating technique ofthe present invention is applied.

[Virtual Segment Interleaving CDMA Transmission System]

In the above, has been described the case in which, in each segmentwithin a symbol, a spread code given by the i-th row of a Walsh functioncorresponds to a code consisting of successive 32 codes starting from0th column of the Walsh function.

This Walsh function having the code length of 32 is written as W₃₂. TheWalsh function W₃₂ has the structure given by the equation 1, 32 codesequences each having the code length of 32 is given by sixteen {W₁₆,W₁₆} and sixteen {W₁₆, {overscore (W)}₁₆}, and there 32 code sequencesare orthogonal to one another. Accordingly, 32 spread code sequencesgenerated by the function {tilde over (W)}₃₂, which is obtained byoffsetting 16 columns of W₃₂, is given by sixteen {W₁₆, W₁₆} and sixteen{{overscore (W)}₁₆, W₁₆}, and the newly generated 32 code sequences areorthogonal to each other. Here, the function {tilde over (W)}₃₂ is givenby the following equation. $\begin{matrix}{W_{2n} = {\begin{matrix}W_{N} & W_{N} \\{\overset{\_}{W}}_{N} & W_{N}\end{matrix}}} & (44)\end{matrix}$

Further, the validity of the orthogonality is not limited to the casethat the offset quantity is 16, and it is easily known that the equation44 is valid even when N is changed from N=16 to 8, 4, 2, 1 successively.Of course, the equation is valid also in the case of increasing N, asN=32, 64, and so on.

This property that orthogonality of the code sequences is alwayssatisfied even when those sequences are offset by any number depends onthe essential property of the Walsh function shown in the equation 1.Utilizing such orthogonality of the Walsh function, it is possible toset a segment that corresponds to a spread code subjected to offset byany quantity. When these offset segments and non-offset segments areintermixed, and it is necessary in particular to distinguish thosesegments, then it is assumed that a “basic segment” means a segmentcorresponding to a spread code starting from the 0th column, and a“virtual segment” means, for example, a segment corresponding to aspread code starting from the 16th column, obtained by offsetting onlythe 16th column.

In the case that optional offset is 16, the relation between the basicsegments and the virtual segments is shown in FIG. 14. In the figure,the horizontal axis indicates chip intervals, and the figure shows thecase in which 128 chip intervals, from 0th to 127th, exist in thecurrent symbol interval. Since the spread code sequence length is set to32, 0th-31st chip intervals constitute 0th basic segment correspondingto 0th-31st spread code sequences. Successive 32nd-63rd chip intervals,64th-95th chip intervals, and 96th-127th chip intervals constitute 1st,2nd, 3rd basic segments, respectively.

On the other hand, 16th-47th chip intervals constitute 1st virtualsegment corresponding to a spread code sequence of 16th-(31st)-15thcolumns having offset 16. Successive 48th-79th chip intervals constitute2nd virtual segment corresponding to a spread code sequence of16th-(31st)-15th columns having offset 16. And, successive 87th-111thchip intervals constitute 3rd virtual segment corresponding to a spreadcode sequence of 16th-(31st)-15th columns having offset 16.

A features of the present invention lies in that despread signal isobtained in both the basic segment and virtual segment. Namely, in theconventional system, the products of the despread code sequence and thedemodulated signal are summed only in the basic segments, and thoseobtained sums are given as the despread signals corresponding to thebasic segment periods. On the other hand, the feature of the presentinvention lies in the improvement that, in addition to the conventionaloperation of summing the products of the despread code and thedemodulated signal in the basic segments, simple operation of summingalso in the virtual segments is carried out, to obtain the despreadsignals corresponding to the basic segment periods and virtual segmentperiods.

When the offset between the basic segment and the virtual segment is 16,duplicate interleave structure is realized as shown in FIG. 14. When theoffset is made smaller than 16 to promote interleave hierarchy,correlation between noises entering the basic segment period and thevirtual segment period is increased to saturate the effect of improvingnoise suppression ability as one feature of the CDMA transmissionsystem, and the improvement effect becomes smaller in comparison withload volume of circuit. Thus, it is not favorable to make the offsetexcessively smaller to promote interleave.

When the offset is set to about half of the spread code length in theWalsh function, for example, offset is set to 16 for the code length of32, then, as described above, interleave becomes the duplicatestructure, and improvement of noise suppression ability is effective. Inthat case, it is possible to obtain communication quality comparable tothe CDMA transmission system in which twice the number of basic segmentsare used for conducting communication.

Namely, communication quality of nearly same level as the case of 8.192cps with 8 basic segments within a symbol is obtained by 4.096 cps withhalf the number of basic segments, i.e. 4 basic segments. In this case,as shown in the figure, four basic segments and three virtual segmentsare used for conducting communication. Despread signals corresponding tobasic segments are regenerated at intervals of the chip period of thespread code sequence length 32, and despread signals corresponding tothe virtual segments of the duplicate interleave structure are obtainedat intermediate times of the basic segments. This corresponds tocommunication at 8.192 cps, i.e., twice the chip rate in the case ofusing only the basic segments. The despread signal corresponding to 0thvirtual segment offset by 16 chip periods from the 0th basic segment isthe despread signal that coincides with the end of the symbol period.Essentially, the end of the symbol period is a time at which informationphase of the primary modulated wave fluctuates drastically, thus givingan unstable value. Thus, it is not necessary to employ such despreadsignal corresponding to the 0th virtual segment. Accordingly, as shownin the figure, seven segments correspond to effective despread signals.

FIG. 15 is a diagram showing an example of a detailed configuration of avirtual segment interleave despreading circuit (deSS-VSI) according tothe present invention. In the figure, the input terminals 2100-2101,spread code sequence input terminal 22 i, multipliers 2102-2103,accumulators 2104-2105, latch registers (REG) 2106-2107, and outputterminals 2108-2109 have respectively the same functions as thecomponents added with corresponding reference numerals in the detailedconfiguration of the conventional despreading circuit (deSS) shown inFIG. 34, and their detailed description will be omitted. Further, theaccumulators 2604-2605 have the same function as the accumulators2104-2105, and their detailed description will be omitted. The selector(SEL) 2610 has the two input terminals F and V, control terminal C, andone input terminal. When a value of binary control signal applied to thecontrol terminal C is 0, the selector selects the value of the inputterminal F, and when the value of the binary control signal applied tothe control terminal C is 1, the selector selects the value of the inputterminal V. The selected value is outputted to the output terminal 2108.Further, the selector (SEL) 2611 has the two input terminals F and V,control terminal C, and one output terminal. When a value of a binarycontrol signal applied to the control terminal C is 0, the selectorselects the value of the input terminal F, and when the value of thebinary control signal applied to the control terminal C is 1, theselector selects the value of the input terminal V. The selected valueis outputted to the output terminal 2109. The demultiplexing circuit(DMPX) 2612 has one input terminal, the two output terminals F and V,and control terminal C. An input signal applied to the input terminal2110 is outputted exclusively to the output terminal F when a binarycontrol signal applied to the control terminal C is 0, or exclusively tothe output terminal V when the binary control signal applied to thecontrol terminal C is 1. The binary counter (BCNT) 2613 is initializedsuch that its output becomes 0 at each point of leading edge of thefirst chip interval of a basic segment, such as the chip intervals 0,128, and so on. Thereafter, the binary counter 2613 counts the number oftimes of applying a signal to the input terminal 2110, and outputs avalue 0 or 1 of the binary least significant digit of the result ofcounting.

Here, the accumulators 2604-2605, selectors (SEL) 2610-2611,demultiplexing circuit (DMPX) 2612, and binary counter (BCNT) 2613 are aseries of components added for obtaining a despread signal correspondingto a virtual segment in the duplicate interleave despreading.

Next, the operation of the virtual segment interleave despreadingcircuit (deSS-VSI) will be described in due order. Corresponding to eachchip, the multiplier 2102 outputs the product of an in-phase componentof the demodulated signal applied to the input terminal 2100 and adespread code sequence applied to the input terminal 22 i. The output ofthe multiplier 2102 is successively inputted to the accumulators 2104and 2604, and accumulated there at the time of trailing edge of a chip.Similarly, corresponding to each chip, the multiplier 2103 outputs theproduct of a quadrature component of the demodulated signal applied tothe input terminal 2101 and the despread code sequence applied to theinput terminal 22 i. The output of the multiplier 2103 is successivelyinputted to the accumulators 2105 and 2605, and accumulated there at thetime of trailing edge of a chip. Here, all the accumulated values arereset to zero at a trailing edge of a reset signal applied to each resetterminal R. In each accumulator to which the reset signal is applied atintermediate times of the first chip intervals of the segments, such aschips 0, 16, 32 and so on, the accumulator is reset at a trailing edgeof the reset signal to have the accumulated value 0. However, just atthe trailing edge of the same chip, an input value and the accumulatedvalue 0 of that accumulator are added and the result of addition is heldby the accumulator. Such operation is repeated by the multipliers2102-2103, accumulators 2104-2105 and 2604-2605, for each chip.

On the other hand, for every 16th chip such as the chips 0, 16, 32, 48and so on, the reset signal is applied to the terminal 2110 at anintermediate time of the first chip interval of each segment. The binarycounter (BCNT), demultiplexing circuit (DMPX), selectors (SEL)2610-2611, and latch registers (REG) 2106-2107 carry out cyclicoperation with respect to a reset signal, as described in the following.

At the leading edge of the first chip in a symbol interval, such as thechips 0, 128, and so on, the binary counter (BCNT) is initialized andits output becomes 0. The binary counter (BCNT) outputs the value 1 atthe time when an odd number of reset signals are inputted, and the value0 at the time when an even number of reset signals are inputted,repeatedly. By this, the binary counter (BCNT) provides alternateoutputs with respect to reset signals, outputting the value 0 always atthe leading edge of the first chip interval of a basic segment, such asthe chips 0, 32, 64, 96, and so on, and outputting the value 1 always atthe leading edge of the first chip interval of a virtual segment, suchas the chips 16, 48, 80, 112, and so on.

Accordingly, when a reset signal is applied in the chip interval 0, thelatch registers (REG) 2106 and 2107 take in and hold the accumulatedvalue of the product of the demodulated signal and the despread codesequence in successive 32 chip intervals before the application of thereset signal, i.e. the chip intervals −32, −31, . . . , −1, from therespective accumulators 2104 and 2105. Those held values are thein-phase and quadrature components of the despread signal in the basicsegment 0. At the same time, the reset signal is led through the binarycounter (BCNT) and demultiplexing circuit (DMPX) to the terminals R ofthe accumulators 2104 and 2105. Awaiting a trailing edge of the resetsignal, the accumulators 2104 and 2105 are reset, and, at the same time,the binary counter (BCNT) is increment. As a result of the stepping ofthe binary counter (BCNT), the output value turns to 1, and thedistributing circuit (DMPX) and selectors (SEL) are each connected tothe side of V, finishing preparation for obtaining the despread signalfor the virtual segment by the reset signal in the chip interval 16.Further, at the trailing edge of the chip 0, each of the accumulators2104 and 2105 is initialized while retaining the input value itself asthe accumulated value, finishing a series of operation in the chipinterval 0.

At each tailing edge of the successive chip intervals 1, 2, . . . , 15,four accumulators 2104-2605 accumulate respective inputs.

When a reset signal is applied in the following chip interval 16, at theleading edge of the reset signal, the latch registers (REG) 2106 and2107 take in and hold the accumulated value of the product of thedemodulated signal and the despread code sequence in the successive 32chip intervals before the application of the reset signal, i.e., thechip intervals −116, −15, . . . , 15, from the respective accumulators2604 and 2605. These held values are the in phase and quadraturecomponents of the despread signal in the virtual segment 0. However, inparticular, the despread signal corresponding to the virtual segment 0in the chip interval 16 extends over the symbol end, and, as describedabove, is not employed for judging the received information. At the sametime, the reset signal is led through the binary counter (BCNT) anddemultiplexing circuit (DMPX) to the terminals R of the accumulators2604 and 2605. Awaiting a trailing edge of the reset signal, theaccumulators 2604 and 2605 are reset, and, at the same time, the binarycounter (BCNT) is stepped. As a result of the stepping of the binarycounter (BCNT), the output value returns to 0, and the demultiplexingcircuit (DMPX) and selectors (SEL) are each connected to the same side Fas the initial state, finishing preparation for obtaining the despreadsignal for the virtual segment by the reset signal in the next chipinterval 32. Further, at the trailing edge of the chip 16, each of theaccumulators 2604 and 2605 is initialized while retaining the inputvalue itself as the accumulated value, finishing a series of operationin the chip interval 16.

Further, at each trailing edge of the successive chip intervals 17, 18,. . . , 31, four accumulators 2104-2605 accumulate respective inputs.

In the chip interval 32, operation similar to the chip interval 0 iscarried out. Similarly, the virtual segment interleaving despreadingcircuit cyclically repeats operation for every 32 chip intervals. Thus,operations in the following chip intervals can be easily inferred, andtheir description will be omitted.

As described above, by using the virtual segment interleavingdespreading circuit (deSS-VSI), it is possible to carry out despreadoperation by interleaving the segments, without adding any processing onthe transmission side of the CDMA transmission system. By substitutingthis virtual segment interleaving despreading circuit (deSS-VSI) foreach of the despreading circuits (deSS) 210-21 n in FIG. 4, it ispossible to realize the virtual segment interleaving despread CDMAtransmission system to which the virtual segment interleave despreadingtechnique of the present invention is applied. Or, in the FIG. 35showing the conventional CDMA transmission system, by substituting thevirtual segment interleaving despreading circuit (deSS-VSI) for each ofthe despreading circuits (deSS) 210-21 n, it is possible to realize thevirtual segment interleaving despreading CDMA transmission system towhich the virtual segment interleaving despreading technique of thepresent invention is applied.

As described above referring to the embodiments, the present inventioncan provide a large-capacity CDMA transmission system that can conductcommunication with a high speed moving unit such as an automobile,transmitting more than same quantity of information as the conventionalsystem using the same frequency band width without deterioratingcommunication quality and without increasing occupied frequency bandwidth in the CDMA transmission system.

FIGS. 16-18 shows the simulation results in the cases that communicationis conducted using the differential CDMA transmission system of thepresent invention, in the same three telephone modes as defined inrelation to the description of FIGS. 36-38. The vertical axes indicateBER and the horizontal axes Eb/No. As the system condition in thesimulations, it is assumed that the transmission frequency domain is 2GHz, chip rate is 4.098 Mcps, information transmission rate is 2.048Mbps, symbol rate is 32 ksps, and spread code sequence length is 32.

When the spread code length is 32 in the conventional CDMA transmissionsystem, it is necessary to assign one channel to the pilot channel, andthus, the number of the information channels is 31 at maximum, and themaximum rate of the information transmission rate is 31×64 kbps=1.984Mbps. On the other hand, in the differential CDMA transmission system, apilot channel is not required, and all channels can be assigned asinformation channels, and the maximum transmission rate is 32×64kbps=2.048 Mbps, which is a first effect.

In the stationary telephone mode (diffCDMA.sty) of the differential CDMAtransmission system, as shown in FIG. 16, the critical transmission bandwidth is 2.56 MHz. Being affected by the large capacity by thetransmission rate of 2.048 Mbps, a floor is generated when the bandwidth is limited to 2.33 MHz or less. However, the frequency utilizationefficiency in this case requires an extremely high state of 0.88 bit/Hzor more. Such highly efficient sophisticated frequency utilization liesin the area that can not be realized without applying the virtualsegment interleave technique disclosed by the present invention. In thecase of the currently utilized CDMA transmission system, the mostsuperior data is the frequency utilization efficiency of about 0.5bit/Hz. Thus, the effectiveness of the present invention is obvious.

Next, in the case that a stationary state of the stationary telephonemode changes to a slowly moving state of 10 km/h of the pedestriantelephone mode (diffCDMA.man), a second effect of the differential CDMAtransmission system can be acknowledged as follows. As shown in FIG. 17,in the differential CDMA transmission system, the critical transmissionband width is observed to be 2.25 MHz. In comparison with the fact that,in the conventional CDMA transmission system, the critical transmissionband width is 3.46 MHz as shown in FIG. 37, the effect of thedifferential CDMA transmission system is remarkable. In the automobiletelephone mode (diffCDMA.car) travelling at further high speed of 100km/h as shown in FIG. 18, the critical transmission band width can beobserved to be 6.74 MHz. As shown in FIG. 41, in the conventional CDMAtransmission system, communication can not be conducted even the widesttransmission band width. In contrast, in the differential CDMAtransmission system, although somewhat strong received electric field ofEb/No≦20 dB is required, high quality communication with BER≦0.001 canbe provided, which clearly shows the dominance of the system.

Thus, the reason that the differential CDMA transmission system has thesuperior transmission characteristics can be understood from the powerspectrum distribution of each channel on the transmission path shown inFIG. 19. In the figure, the vertical axis indicates power, thehorizontal axis indicates frequency, f_(c) indicates carrier frequency,and f_(c)±W/2 indicates upper and lower limit frequencies of thetransmission band.

Namely, the spectrum of the information i channel, shown by the whitearea, and the spectrum of the pilot channel, shown by the shaded area,are different from each other in their spread code sequences used. As aresult, as shown in the figure, they have different frequencycharacteristics from each other. When frequency-selective fading such asmulti-ray Rayleigh fading arises, each channel suffers from distortionhaving the frequency characteristics. This is the phenomenon called thefrequency-selective fading, and this effect becomes stronger as themoving speed increases.

In the case of the pilot in the conventional CDMA transmission system,effects of fading etc. are removed by subtracting the phase errorarising in the pilot channel from the phase error arising in theinformation channel shown in FIG. 19. In that case, when the distortionhaving the frequency characteristics is caused, the error in the pilotchannel and the error in the information channel are different from eachother, and simple operation of obtaining the phase difference can notcorrectly suppress disturbances such as fading.

On the other hand, in the differential CDMA transmission systemdisclosed by the present invention, the fading error is removed bycountervailing phase errors in adjacent symbol intervals. Fortransmitting information, only a specified channel is used. Thus, evenwhen spectrum of a specific channel suffers from distortion havingfrequency characteristics, the spectrum transmitting the informationreceives the same distortion only, which lessening the effect of thefrequency-selective fading. Further, in comparison with the fadingperiod, the symbol interval is much shorter, and thus, the frequencycharacteristics of the frequency-selective fading distortion becomequasi-stationary in adjacent symbol intervals. Thus, by countervailingthe phase errors in the adjacent symbol intervals, the differential CDMAtransmission system can suppress the frequency-selective fading almostcompletely, which can realize a new level of high quality communication.On the other hand, with respect to slight deficiency that appears whenthe transmission band width is set to 6.40 MHz or less in the automobiletelephone mode, it is considered that such deficiency is generated sincethe band width of the transmission signal is diffused over the limitedfrequency band width owing to a high speed fading phenomenon.

FIG. 20 (is95_cp.man) shows of simulating communication in theabove-described pedestrian telephone mode in the case that the phasecontinuous technique of the present invention is applied to theconventional CDMA transmission system.

The vertical axis indicates BER, and the horizontal axis indicatesEb/No. The critical transmission band width is observed to be 3.28 MHz.Thus, as obvious from comparison with the critical transmission bandwith of 3.46 MHz observed in the conventional CDMA transmission systemof FIG. 37 to which the phase continuous technique is not applied, theband width can be narrower. Further, FIG. 21 (diffCDMA_cp.man) showsresults of simulating communication in the same pedestrian telephonemode (diffCDMA_cp.man) in the case that the phase continuous techniqueis applied to the differential CDMA transmission system of FIG. 1. Thecritical transmission band width can be observed to be 2.17 MHz, whichis further narrower than the critical transmission band width 2.25 MHzof the differential CDMA transmission system shown in FIG. 17. Thus, theeffect of the phase continuous technique is confirmed.

FIG. 22 (diffCDMA_cs.man) shows results of simulating communication inthe pedestrian telephone mode in the case that the chip waveformcontinuating technique is applied to the differential CDMA transmissionsystem of FIG. 1. The vertical axis indicates BER, and the horizontalaxis indicates Eb/No. The critical transmission band width is observedto be 2.17 MHz, which is narrower than the critical transmission bandwidth 2.25 MHz of the differential CDMA transmission system of FIG. 17to which the chip waveform continuating technique is not applied. Thus,the effect of the chip waveform continuating technique is confirmed.

FIG. 23 (is95_CPS.man) shows results of simulating communication in thepedestrian telephone mode in the case that the phase continuoustechnique and chip waveform technique of the present invention areapplied to the conventional CDMA transmission system. The vertical axisindicates BER, and the horizontal axis indicates Eb/No. The criticaltransmission band width is observed to be 3.28 MHz, which is narrowerthan the critical transmission band width 3.46 MHz in the conventionalCDMA transmission system of FIG. 37 to which the phase continuous andchip waveform continuating techniques are not applied. Further, FIG. 24(diffCDMA_cps.man) shows results of simulating the pedestrian telephonemode (diffCDMA_cps.man) in the case that the phase continuous techniqueand chip waveform continuating technique are applied to the differentialCDMA transmission system shown in FIG. 1. The vertical axis indicatesBER, and the horizontal axis indicates Eb/No. The critical transmissionband width is observed to be 2.14 MHz, which is narrower than thecritical transmission band width 2.25 MHz in the conventional CDMAtransmission system of FIG. 37 to which the phase continuous techniqueand chip wave continuating technique are not applied. Thus, the effectof simultaneous applying the phase continuous technique and chipwaveform continuating techniqe is clearly shown.

FIG. 25 (diffCDMA_VSI.car) shows result of simulating communication inthe automobile telephone mode in the case that the virtual segmentinterleave technique is applied to the differential CDMA transmissionsystem of FIG. 1. The vertical axis indicates BER, and the horizontalaxis indicates Eb/No. The critical transmission band width is observedto be 0.80 MHz, which is narrower than the critical transmission bandwidth 6.74 MHz in the differential CDMA transmission system of FIG. 18to which the chip waveform continuating technique is not applied. Thus,the effect of the virtual segment interleaving technique can beconfirmed. In this critical transmission band width, the frequencyutilization efficiency is 2.5 bit/Hz, and showing remarkable improvementeffect.

In the above-described embodiments, the present invention has beendescribed taking an example of the radio transmission system. However,the present invention can be applied to the optical communication systemusing fiber etc., and similar effects as in the application to the radiocommunication can be obtained. Namely, drift of an emission wavelength,i.e. oscillating frequency of laser usually used as a light source whichis a disturbance factor in the optical communication system, can be madeto correspond to the already-described Doppler shift in the radiotransmission system. As a mode of light that proceeds through a fibercore, may be mentioned a mode in which light proceeds straightly, and amode in which light proceeds being slightly inclined and repeatingreflection between a core and clad, for example. Lights of those modesinterfere with each other, and this interference can be made tocorrespond to the fading phenomenon in the radio transmission system.Further, when optical communication develops from intensity to multiplecoherent communication, drift of the oscillating frequency of a lightsource corresponding to the Doppler shift, inter-mode interference andinter-wavelength interference corresponding to the multi-ray fading willbecome more obvious, and the effect of the present invention will bemore remarkable.

1. A code division multiple access transmission system, comprising: on atransmitting side, a means for obtaining a primary modulated wave byperforming differential coding phase modulation on a carrier signal inaccordance with information; and a means for generating a spread signalincluding a plurality of basic and virtual segments, by multiplying saidprimary modulated wave by a spread code repeatedly a plurality of times,changing a time region, via said basic and virtual segments, within asymbol period, and for transmitting said generated spread signal; and ona receiving side, a means for detecting a phase difference between apast symbol and a present symbol, by performing quasi-synchronousdetection and despreading, and difference operation; and a means foroutputting the detected phase difference as information of said symbol.2. A code division multiple access transmission system, comprising: on atransmission side, a means for obtaining a primary modulated wave byperforming phase modulation on a carrier signal in accordance withinformation; a means for excluding rapid fluctuation of a phase value ina symbol end area of said primary modulated wave; and a means forgenerating a spread signal by multiplying said primary modulated wave,from which the rapid fluctuation of the phase value is excluded, by aspread code, and for transmitting said generated spread signal; and on areceiving side, a means for regenerating the information by despreading,said despreading being performed by obtaining a sum of values that, inturn, are obtained by multiplying the received spread signal by acorresponding despread code.
 3. A code division multiple accesstransmission system, comprising: on a transmitting side, a means forobtaining a primary modulated wave by performing phase modulation on acarrier signal in accordance with information; a means for excludingrapid fluctuation of a value of a spread code in an end area of a spreadcode period; and a means for generating a spread signal by multiplyingsaid primary modulated wave by a spread code, from which the rapidfluctuation of the value of the spread code is excluded, and fortransmitting said generated spread signal; and on a receiving side, ameans for regenerating the information by despreading, said despreadingbeing performed by obtaining a sum of values that, in turn, are obtainedby multiplying the received spread signal by a corresponding despreadcode.
 4. A code division multiple access transmission system,comprising: on a transmitting side, a means for obtaining a primarymodulated wave by performing phase modulation on a carrier signal inaccordance with information; and a means for generating a spread signalincluding a plurality of basic and virtual segments, by multiplying saidprimary modulated wave by a spread code sequence repeatedly a pluralityof times, via said basic and virtual segments within a symbol period,and for transmitting said spread signal; and on a receiving side, ameans for regenerating the information by despreading, said despreadingbeing performed by obtaining a sum of values that, in turn, are obtainedby multiplying basic and virtual segments of a received spread signal bya corresponding despread code sequence; wherein said means forregenerating, on the receiving side, performs said despreading invirtual segments defined by superposing the said basic and virtualsegments, changing a time region.
 5. A code division multiple accesstransmission system comprising: on a transmitting side, a means forobtaining a primary modulated wave by performing differential codingphase modulation on a carrier signal in accordance with information; ameans for excluding rapid fluctuation of a phase value in a symbol endarea of said primary modulated wave; and a means for generating a spreadsignal including a plurality of transmission segments, by multiplyingsaid primary modulated wave by a spread code repeatedly a plurality oftimes, changing a time region within a symbol period, and fortransmitting said spread signal; and on a receiving side, a means fordetecting a phase difference between a past symbol and a present symbol,by performing quasi-synchronous detection and despreading, anddifference operation of a received spread signal; and a means foroutputting the detected phase difference as information of said symbol.6. A code division multiple access transmission system comprising: on atransmitting side, a means for obtaining a primary modulated wave byperforming differential coding phase modulation on a carrier signal inaccordance with information; a means for excluding rapid fluctuation ofa spread code in an end area of a spread code period of said spreadcode; and a means for generating a spread signal including a pluralityof transmission segments, by multiplying said primary modulated wave bya spread code repeatedly a plurality of times, changing a time regionwithin a symbol period, and for transmitting said spread signal; and ona receiving side, a means for detecting a phase difference between apast symbol and a present symbol, by performing quasi-synchronousdetection and despreading, and difference operation of a received spreadsignal; and a means for outputting the detected phase difference asinformation of said symbol.
 7. The code division multiple accesstransmission system according to claim 1, further comprising, on thereceiving side: a means for regenerating the information by despreading,said despreading being performed by obtaining a sum of values that, inturn, are obtained by multiplying basic and virtual segments of thereceived spread signal by a corresponding despread code; wherein saidmeans for regenerating, on the receiving sides, performs saiddespreading in virtual segments defined by superposing the said basicand virtual segments, changing a time region.
 8. The code divisionmultiple access transmission system according to claim 5, furthercomprising, on the transmitting side: a means for excluding rapidfluctuation of a spread code in an end area of a spread code period ofsaid spread code.
 9. The code division multiple access transmissionsystem according to claim 5, further comprising, on the receiving side:a means for regenerating the information by despreading, saiddespreading being performed by obtaining a sum of values that, in turn,are obtained by multiplying transmission segments of the received spreadsignal by a corresponding despread code; wherein said means forregenerating, on the receiving side, performs said despreading invirtual segments defined by superposing the transmission segments,changing a time region.
 10. The code division multiple accesstransmission system according to claim 6, further comprising, on thereceiving side: a means for regenerating the information by despreading,said despreading being performed by obtaining a sum of values that, inturn, are obtained by multiplying transmission segments of the receivedspread signal by a corresponding despread code; wherein said means forregenerating, on the receiving side, performs said despreading invirtual segments defined by superposing the transmission segments,changing a time region.
 11. The code division multiple accesstransmission system according to claim 8, further comprising, on thereceiving side: a means for regenerating the information by despreading,said despreading being performed by obtaining a sum of values that, inturn, are obtained by multiplying transmission segments of the receivedspread signal by a corresponding despread code; wherein said means forregenerating, on the receiving side, performs said despreading invirtual segments defined by superposing the transmission segments,changing a time region.
 12. The code division multiple accesstransmission system according to claim 2, further comprising, on thetransmitting side: a means for excluding rapid fluctuation of a spreadcode in an end area of a spread code period of said spread code.
 13. Thecode division multiple access transmission system according to claim 2,wherein: said means for regenerating, on the receiving side, performssaid despreading in virtual segments defined by superposing thetransmission segments.
 14. The code division multiple accesstransmission system according to claim 12, wherein: said means forregenerating, on the receiving side, performs said despreading invirtual segments defined by superposing the transmission segments. 15.The code division multiple access receiving system according to claim 3,wherein: said means for regenerating, on the receiving side, performssaid despreading in virtual segments defined by superposing thetransmission segments.
 16. The code division multiple accesstransmission system according to claim 4, further comprising, on thetransmitting side: a means for excluding rapid fluctuation of a spreadcode in an end area of a spread code period of said spread codesequence.
 17. The code division multiple access transmission systemaccording to claim 7, further comprising, on the transmitting side: ameans for excluding rapid fluctuation of a spread code in an end area ofa spread code period of said spread code sequence.